## 000802 1500802

4.0001 8.000e-1.200e-1.6001 2.000e- .4001 6.588-04-7.941-12 5.858-12 6.318-12 6.651-12 6.868-12 6.961-12 6.938-12 6.778-12 6.498-12 6.098-12-5.608-12 7.198-06 1.768-03 4.021-03 S.131-03 7.918-03 9.228-03 9.988-03 1.018-02 9.678-03-b.628-03 7.068-03 5.108-03 2.908-03 6.588-04

## 0037

Thus for this case, a 10 percent power-supply voltage change results in a 0.37 percent change in out- The result is significantly better than for a bipolar Widlar current source. The MOS counterpart of the base-emitter reference is shown in Fig. 4.366. Here The case of primary interest is when the overdrive of T is small compared to the threshold voltage. This case can be achieved in practice by choosing sufficiently low input current and large (W L)i. In this case, the output current is...

## 004

Figure 2.60 Typical experimental variation of drain current as a function of the square root of gate-source voltage in the active region. Table 2.4 Summary of Process Parameters for a Typical Silicon-Gate w-Well CMOS Process with 0.4 jum Minimum Allowed Gate Length Table 2.4 Summary of Process Parameters for a Typical Silicon-Gate w-Well CMOS Process with 0.4 jum Minimum Allowed Gate Length

## 0045

These circuits are not fully supply independent because the base-emitter or gate-source voltages of T change slightly with power-supply voltage. This change occurs because the collector or drain current of T is approximately proportional to the supply voltage. The resulting supply sensitivity is often a problem in bias circuits whose input current is derived from a resistor connected to the supply terminal, since this configuration causes the currents in some portion of the circuit to change...

## 02

Figure 4.59 Cascode active-load circuit for Problem 4.16. Figure 4.59 Cascode active-load circuit for Problem 4.16. drwn 1 ixm and Xd 0. Use Table 2.3 for other parameters. 4.15 Repeat Problem 4.14, but now assuming that 2 kil resistors are inserted in series with the sources of M3 and M4. Ignore the body effect. 4.16 Determine the unloaded voltage gain v0 v, and output resistance for the circuit of Fig. 4.59. Neglect Verify with SPICE and also use SPICE to plot the large-signal Vq-Vi transfer...

## 03

04 06 > 04 13 13 13 13 13 14 2 mic 2 cmosn 2 m2c 2 cmosn 04 1.023e-04 15 -4.594e-15 14 -1.1121-14 01 7.754e-01 01 6.5231-01 01 -4.5941-01 01 6.656b-01 9.5631-02 1.912e-02 5.2181-01 1.820e-03 3.0761-05 4.081e-04 5.1721-14 1.592e-13 1.716e-13 6.4531-14 1.3661-13 2.240e-14 xcm 2 m3 2 cm0sp -1.0231-04 1.548e-20 2.1881-14 -9.912e-01 -2.1881+00 0. -7.733e-01 -1.875e-01 4.9781-03 4.909e-01 9.2251-04 1.936e-06 2.6711-04 9.103e-14 4.0461-13 4.639e-13 1.511e-13 3.7061-13 3.360e-14 -1.0231-04 -1.000e-04...

## 035

Per unit gate width Threshold adjust implant (box dist) impurity type effective depth effective surface concentration Nominal threshold voltage Polysilicon gate doping concentration Poly gate sheet resistance Source, drain-bulk junction capacitances (zero bias) Source, drain-bulk junction capacitance grading coefficient Source, drain periphery capacitance (zero bias) Source, drain periphery capacitance grading coefficient Source, drain junction built-in potential

## 04

Figure 2.32 Typical device parameters for bipolar transistors in a low-voltage, oxide-isolated, ion-implanted process. Figure 2.32 Typical device parameters for bipolar transistors in a low-voltage, oxide-isolated, ion-implanted process. limitation on the performance attainable in analog circuits, leading to the development of several pnp transistor structures that are compatible with the standard IC fabrication process. Because these devices utilize the lightly doped -type epitaxial material...

## 05

Figure 1.45 Transconductance-to-current ratio versus overdrive. the remainder of this book and assume that MOS transistors operate in weak inversion for overdrives less than the bound given in (1.255). Equation 1.208 can be used to find the transition frequency. In weak inversion, Cgs Cgd 0 because the inversion layer contains little charge.39 However, Cgh can be thought of as the series combination of the oxide and depletion capacitors. Therefore, Substituting (1.253) and (1.256) into (1.208)...

## 082

Ig myv o- roz 1 2mOr0. + r*l 4 CMOSH K-IOOO L 1U 0 CMOSH W 1000 L 10 BOTE TEAT CONNECTING THE BODY 10 THE SOCRCE ELIMINATES .MODEL CMOSH WOS LEVEL 1 LAKBDA 0.024 VTO-O.6 KP-1940 LD 0.09D DOTE THAT LAMBDA * (DZD DVDS) LEFF 0.02 0.82 0.0244 OPTIONS HOHOD NO AGE .WIDTH OOTB80 DC TRAHSFER CORVES ITCH- 27.000 TEMP- 27.000

## 082 082

This calculation assumes that dXd dVDS and Leff are constant for each type of transistor, allowing us to use constant Early voltages. In practice, however, dXd dVDS and Leff both depend on the operating point, and accurate values of the Early voltages are rarely available to circuit designers when channel lengths are less than about 1.5 xm. As a result, circuit simulations are an important part of the design process. SPICE simulation of the op amp under the conditions described above gives a...

## 09

Compare your answer with a SPICE simulation. Also, compare your answer to the result that would apply without mismatch. 4.19 Although Gm cni of a differential pair with a current-mirror load can be calculated exactly from a small-signal diagram where mismatch is allowed, the calculation is complicated because the mismatch terms interact, and the results are difficult to interpret. In practice, the mismatch terms are often a small fraction of the corresponding average values, and the...

## 0q10

0 hph 4.048e-08 1.0121-05 5.542e-01 1.287e+00 -7.332e-01 1.3718+01 1.305e-05 2.500e+02 3.912e-04 6.390e+05 0. 0 014 0 php -9.7308-08 -4.8658-06 -5.590e-01 -5.5908-01 0. 1.3718+01 2.7748-06 5.000e+01 1.8818-04 2.658e+05 0. 0 Q11 0 hph 4.0488-08 1.012e-05 5.542e-01 5.5421-01 0. 1.4448+01 5.6318-06 2.5008+02 3.9128-04 6.3908+05 0. TS0M-i phase vout 8.8738-23 8.8731-22 the transcohdcfctahce cas also b8 measured by 8luchatibs the voltage 30urc8 comected at toe cctput and the ac analysis above,...

## 1

In this equation, Vr is the reverse bias on the diode and n has a value between 3 and 6. The operation of a pn junction in the breakdown region is not inherently destructive. However, the avalanche current flow must be limited by external resistors in order to prevent excessive power dissipation from occurring at the junction and causing damage to the device. Diodes operated in the avalanche region are widely used as voltage references and are called Zener diodes. There is another, related...

## 1 1

Equation 3.84 shows that the body effect reduces the output resistance, which is desirable because the source follower produces a voltage output. This beneficial effect stems from the nonzero small-signal current conducted by the gmb generator in Fig. 3.25b, which increases the output current for a given change in the output voltage. As ra oo and Rl oo, this output resistance approaches l (gm + gmb). The common-gate input resistance given in (3.54) approaches the same limiting value. As with...

## 1 1 1

Figure 9.45 Poles of the transfer function of the feedback amplifier of Fig. 9.41. The transfer function contains no zeros. near the jco axis, which would give rise to an excessively peaked response. In practice, oscillation can occur because higher magnitude poles do exist and these would tend to give a locus of the kind of Fig. 9.39, where the remote poles cause the locus to bend and enter the right half-plane. (Note that this behavior is consistent with the alternative approach of...

## 1 1 1 jsiIsO JVApF49

The first term in (4.9) stems from finite output resistance and the second term from finite If Vce2 > Vcei the polarities of the two terms are opposite. Since the two terms are independent, however, cancellation is unlikely in practice. The first term dominates when the difference in the collector-emitter voltages and 3p are large. For example, with identical transistors and VA 130 V, if the collector-emitter voltage of Q is held at vbe (on), and if the collector-emitter voltage of q2 is 30...

## 10

Allowed dimensions of passive devices have also decreased.) This trend is driven primarily by economics in that reducing dimensions increases the number of devices and circuits that can be processed at one time on a given wafer. A second benefit has been that the frequency capability of the active devices continues to increase, as intrinsic fj values increase with smaller dimensions while parasitic capacitances decrease. Vertical dimensions such as the base width of a bipolar transistor in...

## 10 V

Figure 11.54 Differential-pair input stage for Problem 11.14. figure of this circuit in decibels for 10 kft using the following data. Neglect flicker noise and capacitive effects. (b) If the device has fT 500 MHz, calculate the frequency where the noise figure is 3 dB above its low-frequency value. 11.20 (a) Neglecting capacitive effects, calculate the noise figure in decibels of the circuit of Fig. 11.52 with Rs 5 kft. Use data as in Problem 11.10. (b) If the flicker noise corner frequency for...

## 100

Hi9 Ids - Ic23 (0.22 - 0.05) mA 0.17 mA Substitution in (5.84) and rearranging gives Vbei9 Vbe 18 -Vbe23 (26 mV) In 613 mV Vbe2Q (Vbel9 + Vben - VbeU) -525 mV l& o -5.9 (JLA and the collector current in Q20 is quite small as predicted. Finally Icl7 0.68 mA - fo.68 - mA 0.68 mA

## 100 1 1 2 2 2

. deel hpv hph bmoooo w 130 . opticus nomod hopage I I -2.SOCI-03 -2. 001-03 - 2.100E-03 -2.2001-03 -2.00CE-03 1.DOOMO 2-OOOE+OO 3.0002+00 1.0008.50 5-300E+00 6.000E 00 7.0001+00 8.OOOE+OO 9.OOOE+OO 1.000E+01 1.100E+01 1.200E+01 1.3 00E+01 l.lOOE+Ol 1.5001+01 1.600E+01 1.700 -01 . OOE-Ol 1.> 001-01 2-JOOE'Ol 2.100E-01 2.200E-01 2.3001+01 2.1001*01 2.500E+01 2.600E+01 2.7001+01 2.IOOE.C1 -2.111-03-* -2.151-03 < -2.171-03 < -2.1 -03 -2.20E-03 . -2.221-03 ' -2.241-03 + -2.251-03 -2.211-03...

## 100 200 300 400

0 Q1 0 H 1.009e-06 1.0098-04 7.744e-01 3.0978+00 -2.322e+00 -3.097e+00 3.131e-04 1.000e+02 3.899e-03 2.564e+04 0. 0 q5 0 h 1.000e-06 1.0001-04 7.742e-01 7.742e-01 0. -2.322e+00 7.8201-05 1.000e+02 3.867e-03 2.586e+04 0. 0 Q10 0 h 9.769e-07 9.769e-05 7.736e-01 7.736e-01 0. -7.736e-01 7.633e-05 1.000e+02 3.777e-03 2.647e+04 o. -1.029e-06 -1.029e-04 -7.749e-01 -1.799e+00 1.024e+00 -4.121k+00 1.8598-04 1.000K+02 3.978e-03 2.514e+04 0. 0 Q6 0 m 1.000e-06 1.000e-04 7.742e-01 7.742e-01 0. -1.548e+00...

## 1000

This gain is 0.3 percent more than the quiescent value. At the negative signal peak, the transistor collector current is IQ- Io 1.86 - 0.6 1.26 mA 20.6 fl and use of (5.22) gives the small-signal gain as 1000 This gain is 0.7 percent less than the quiescent value. Although the collector-current signal amplitude is one-third of the bias current in this example, the small-signal gain variation is extremely small. This circuit thus has a high degree of linearity. Since the nonlinearity is small,...

## 1000102

*** operatins poiht information ts3m 27.000 tekp 27.000 bode -voltage noes voltage node -voltage 1.594e+00 0 3 0. 0 6 - 2.398e+00 0 100 I.OOOE-1.000 -6.0002-8.0002-1.000B- 1.53E+00-+-1.598+00 1.59E-+00 1.59E+0Q 1.592+00 1.55E+00 + 1.592+00 1.552+00 1.551+00 1.592+00 + 1.59E+00-+-- -voltage 7.986e-01 - 2.398e+00 --2.500e+00 * * mosfets subcat ktikmwtt model

## 1000104 2000104 3000804 4000804

TRANSIBtf ANALYSIS TB*. 27.000 TEUF 27.000 It ) -4.0002*00 -2.OOOE+OO 0. 2.0001*00 (.0001*00 0. 4.0001-06 4.SIE-Ol . . A . 8.0001-06 9.15E-01 + i . 1.2008-05 1.328+00 + + > ' 1 + < + 1.6008-05 1.6(8+00 + + * + 2.0008-05 1.86E+00 A . . 3.2001-05 1.76E+00 t + A + + . 3.600E-05 l.(9E+00 . + + . * + 4.0001-05 4.4001-05 6.B61-01 + + . A . + + 4.1008-05 2.151-01 . > 1 ' + 5.2001-05 -2.B21-01 < A + 5.6001-05 -7.748-01 . + * A . . . 6.0001-05 -1.221+00 + + 1 . + + + + 6.4008-05 -1.591+00 + A *...

## 1000e03 1500k03

250E-.500 .750E-OOOE .250E--5C0E OOOE-.25DE -500E .'501-. 000E .I50E--503E-.7501- 0C0E-.250E-.500E-.75 . 0001-253E-5001-.50E-I 1 .S2E-04-' 1.82E-04 1.B2E-C4 1.B2E-04 ' 1.B2E-04 B.21E-0I < S.24E-CI 8.256-04 9.2 E-04 i S.2TE-0I . 3.2BE-C4-' .29E-04 B.31E-C4 B.32E-04 B.33E-04 B.3IE-04 . B.35E-04 * B.36E- 4 * B.3E-si f B.3BE-01 S.I0E-C4 S.I2E-CI S.43E-0I .iie i S.45E-04 B.liE-0 e.CE-34 B ISE-Ot S.491-11 MODEL PHP PHP IS 1E-16A BF-50 OPTIONS NOPAGS NONOD .WIDTH OOT-80 .OPTIONS SPICE

## 101

3.3.7 Common-Drain Configuration (Source Follower) The common-drain configuration is shown in Fig. 3.25a. The input signal is applied to the gate and the output is taken from the source. From a large-signal standpoint, the output voltage is equal to the input voltage minus the gate-source voltage. The gate-source voltage consists of two parts the threshold and the overdrive. If both parts are constant, the resulting output voltage is simply offset from the input, and the small-signal gain would...

## 101 Introduction

Chapters 1 through 9 have dealt almost entirely with analog circuits whose primary function is linear amplification of signals. Although some of the circuits discussed (such as Class AB output stages) were actually nonlinear in their operation, the operations performed on the signal passing through the amplifier were well approximated by linear relations. Nonlinear operations on continuous-valued analog signals are often required in instrumentation, communication, and control-system design....

## 102

Khlbtear fonctioh stothesis, add rs akd re 912 13 14 16 p rs12 16 1 ix qll 0 15 14 p reias 15 0 33.4x 96 3 3 2 r ii 2 o 100va 9715(1 98 6 6 7 8 915 7 10 11 r rq15 11 o ix 89 8 7 9 h 914 1 13 10 i v2vbe 20 o 1.56v vdomft 20 8 ov nddel i hph bfaloo is 1e-17 rb-200 ri2 .h3del p pnp bf 100 is-1e-17 rb-200 re-2 dc ii O 500U ion plot dc i(vro nr) .opticrs ndpagi n08bd '.width cct 80 .orneas spice

## 102 Precision Rectification

Perhaps the most basic nonlinear operation performed on time-varying signals is rectification. An ideal half-wave rectifier is a circuit that passes signal currents or voltages of only one polarity while blocking signal voltages or currents of the other polarity. The transfer characteristic of an ideal half-wave rectifier is shown in Fig. 10.1. Also shown in Fig. 10.1 is the transfer characteristic of a second useful rectifier, the full-wave type. Practical rectifiers can be divided into two...

## 1034 A Complete Analog Multiplier3

In order to be useful in a wide variety of applications, the multiplier circuit must develop an output voltage that is referenced to ground and can take on both positive and negative values. The transistors Q3, Q4, Q5, Q6, Q7t and 8, shown in Fig. 10.13, are referred to as the multiplier core and produce a differential current output that then must be amplified, converted to a single-ended signal, and referenced to ground. An output amplifier is thus required, and the complete multiplier...

## 1035 The Gilbert Multiplier Cell as a Balanced Modulator and Phase Detector

The four-quadrant multiplier just described is an example of an application of the multiplier cell in which all the devices remain in the active region during normal operation. Used in this way the circuit is capable of performing precise multiplication of one continuously varying analog signal by another. In communications systems, however, the need frequently arises for the multiplication of a continuously varying signal by a square wave. This is easily accomplished with the multiplier...

## 104 Phase Locked Loops PLL

The phase-locked loop concept was first developed in the 1930s.4 It has since been used in communications systems of many types, particularly in satellite communications systems. Until recently, however, phase-locked systems have been too complex and costly for use in most consumer and industrial systems, where performance requirements are more modest and other approaches are more economical. The PLL is particularly amenable to monolithic construction, however, and integrated-circuit...

## 1041 Phase Locked Loop Concepts

A block diagram of the basic phase-locked loop system is shown in Fig. 10.18. The elements of the system are a phase comparator, a loop filter, an amplifier, and a voltage-controlled oscillator. The voltage-controlled oscillator, or VCO, is simply an oscillator whose frequency is proportional to an externally applied voltage. When the loop is locked on an incoming periodic signal, the VCO frequency is exactly equal to that of the incoming signal. The phase detector produces a dc or...

## 1042 The Phase Locked Loop in the Locked Condition

Under locked conditions, a linear relationship exists between the output voltage of the phase detector and the phase difference between the VCO and the incoming signal. This fact allows the loop to be analyzed using standard linear feedback concepts when in the locked condition. A block diagram representation of the system in this mode is shown in Fig. 10.20. The gain of the phase comparator is KD V rad of phase difference, the loopfilter transfer function is F(s), and any gain in the forward...

## 11

Note that both (1.21) and (1.22) predict values of Cj approaching infinity as Vd approaches i o- However, the current flow in the diode is then appreciable and the equations no longer valid. A more exact analysis2 3 of the behavior of Cj as a function of Vd gives the result shown in Fig. 1.3. For forward bias voltages up to about i y0 2, the values of C predicted by (1.21) are very close to the more accurate value. As an approximation, some computer programs approximate Cj for VD > < 0 2 by...

## 110

Hph bf> to is-12-18 php bf 20 is-1e-18 . del pmds1 pm08 hp 2so vto--0.7 lambda-0 ld 0 model p s2 phos kp 2sc vto -0.7 lambda*0 lb 0 the dc ihput voltage is adjusted bt trial ahd error to set the dc output voltage to zero. opncss mdpags bcmod . fli ih (x7t 80 BICM08 CLASS-AB output stage (peak output amplitude 2 V) bipolar parameters prom fig. 2.32 add pics parameters from table 2.3

## 111

Figure 10.15 Input and output spectra for a balanced modulator. This dc component can be introduced intentionally to provide conventional amplitude modulation or it can be the result of offset voltages in the devices within the modulator, which results in undesired carrier feedthrough in suppressed-carrier modulators. Note that the balanced modulator actually performs a frequency translation. Information contained in the modulating signal Vm(t) was originally concentrated at the modulating...

## 1115

Check headroom requirements in the bias circuit For the. branch induct ing M,3 0, this branch operate with all transistors in the active region if Voy C 0-33 V For the branch including M 0.1 v) - VT in 4- 0.7 v- (-ifc). 0 U,. in 4 36 mV (neglect) - -0.8 V 5 -0. -0 -0 -0.7-(-l'5) 3 y 1.3 3 V*, o. 43 V So, this branch operates with all -transistors in the active, region if H < 0.43 V 50 this lesion vvill u e the previous y cormputeef value .of

## 116

Thus, for stability, the Nyquist criterion requires that T0 < 11.6 and this is close to the answer obtained from the root locus. If the point on the jw axis where the root locus crossed had been determined more accurately, it would have been found to be at 3.8 x 106 rad s and both methods would predict instability for T0 > 11.6. It should be pointed out that the root locus for Fig. 9.39 shows the movement of the poles of the feedback amplifier as T0 changes. The theory developed in Section...

## 12

To find em, we will again refer to Fig. 4.266 with v,i vt2 vic. In writing (4.134), we assumed that l gm3. We will now reconsider this assumption and write We will still assume that the two transistors in the differential pair match perfectly and operate with equal dc currents, as do the two transistors in the active load. Then (4.174) can be rewritten as Mmir) + 2i0(m(> ) + gm mir) rrr(mir)fo(mir) Substituting (4.175) into (4.135) gives rTr(mir) I 2r0(mir) gm mir)rir(mir)roimir) Substituting...

## 12 V

Figure 5.42 k -npn Darlington output stage. (c) Calculate the maximum average power that can be delivered to RL 8 ft before clipping occurs and the corresponding efficiency of the complete circuit. Also calculate the maximum instantaneous power dissipated in each output transistor. Assume that feedback is used around the circuit so that V0 is approximately sinusoidal. (d) Use SPICE to plot the dc transfer characteristic from Vj to V as V0 is varied over the complete output voltage range with RL...

## 120

In this example, acmc is much larger than acm because the transconductance in (12.35) is much larger than the degenerated transconductance in (12.37). The CMFB loop uses negative feedback to make Voc VCM. If VCM changes by a small amount from its design value due to parameter variations in the circuit that generates Vcm, Kr Should change by an equal amount so that Voc tracks VCM. The ratio A Voc A VCM is the closed-loop small-signal gain of the CMFB loop, which from Fig. 12.12 is 1, Acmfb 88 1...

## 1252114 2661114 3302114 1999114 2302114 3061115

SMALL-SIGHAL TRANSFER CHARACTERISTICS V(16) VII OUTPUT RESISTABCE AT V 16) K ASALTSIS TIB) VI1S Il I 0. 1.00(1-00 2.000fr .000 37 .0591.07 -122E+07 -188E+07 -258E.07 -333E.07 .4128*07 4968-07 -SB4E.C7 .6798*07 7788.07 5838*07 9958*07 1138*07 .2388*07 J718.C7 .5118*07 .6608*07 .8188*07 .9858*07 .1628*0'' .3498*07 .5488*07 .588*07 .9818*07 .2178*07 -166E*C7 .7318*07 .0118*07 -30BE+07 6238*07 9568*07 .3058*07 6838*07 0758*07 49 E.07 94 E 7 4148*07 9128*07 l f.C7 0008*08 838*00-1 518*00 .418*00...

## 1252114 2661114 3302e14 1999114 2302e14 3061e15

1.4921-22 2.378e-20 2.302e-20 6.126e-22 2.302s-20 1.492s-22 1.2521-14 2.661s-14 3.3021-14 1.9991-14 2.302e-14 3.0611-15 1.492e-22 2.378e-20 2.302e-20 6.126e-22 2.302e-20 1.492e-22 1.252e-14 2.661e-14 3.302e-14 1.999e-14 2.3021-14 3.0612-15 0 sjbop 1.000e-04 -1.277e-14 -2.967e-14 1.277e+00 1.690e+00 -1.277e+00 7.000e-01 5.774e-01 6.000e-04 0. 7.7b1e-23 2.402e-20 2.302e-20 9.206e-22 2.302e-20 7.781e-23 1.176e-22 4.800e-20 4.604e-20 1.841e-21 4.604e-20 1.176e-22 1.000e-04 -1.277e-14 -2.967e-14...

## 1258806 1511806 1395806

2.5118.06 3.162E.06 3.5618-06 5.0118 6 6.309E.Q6 7.S13E.36 1.0C0E+07 1.2558+07 1.5318+07 1.395E+07 2.5118*07 3 .1528*07 3.9318-0* 5.0118.07 5.3D98+07 7. 138.07 1.0008*08 1.79E+02-* 1.798+02 < 1.79E+02 1.79E+02 . 1.79E*02 1.798*02 * 1.78E.02 1.788*02 1.78 +02 1.778*02 * 1.778*02-* 1.7 S 02 1.75S.C 1.7 *o * 1.728.02 1.718.02 1.688.02 > 1-65E.02 -1.61 02 1.568.02 + 1. 81*02-* 1.338+02 1.258.02 1.088+02 9.178+01 7.5E+01 6.798*01 6.25E+01 6.00E.01 5.952*01 6.01E+01-* S.OSE.Ol 4.112+01 . 4.I E OI...

## 1280 V

To reduce the TCF, the constant M in (4.249)-(4.252) is often trimmed at one temperature so that the band-gap output is set to a desired target voltage.19 In principle, the target voltage is given by (4.253). In practice, however, significant inaccuracy in (4.253) stems from an approximation in (4.244).20 As a result, the target voltage is usually determined experimentally by measuring the TCp directly for several samples of each band-gap reference in a given process.2122 This procedure reduces...

## 130

Fi-om Curvd A > A0 38,000 mil* eu ne B, A0 - 20,000 um C ijooo mit2- curve a,& j and d axe predicied by t-he quation. 5 nversdy proporfi'onal -to fhe 5 3 , N X 5 tbtf. propor-tionAlihj constant related +0 the i afer siy mort Spea'f cAlly, K 5 effective, or uiablr txrea on tant vtafer). By (2-S6) , -the cost per un if silicon h -tota thermal r sistante 19 att

## 1300106

1.55 i-c6 1-60cs-06 1.6501-05 1.7ccs-35 1.7501-06 1-B002-06 -5.00e*00--4.522*00 -4.042+00 -3.572*00 -3.092*00 -2.612*00 -2.142*00 -1.182*00 -7.081-01 -2.312-01 2.462-01 7.23e-01 2.15E*00 2.63e 00 3.102*03 3.5se.oo 4. ee+g0 4.53e-00 5.ooe o 5.00e*00 SLEW RATE 10V 1.20S 8.33V DS 8.33 1.9 4.4 IMPROVEMENT To ffnd point 0-f SUw limiting put L.2XID4 SI*uJ YfliU. - ew rate Cw roves ttm s (A) coVm

## 131

CK 1.449E-13 1.70 E-13 1.7068-13 1.449E-13 CCS 5.7951-13 7.359E-13 7.3591-13 5.795E-13 BETMC 2.000E+02 2.000E+02 2.000E+02 2.000E+02 PT 7.747E+06 2.071E+08 2.071E+08 7.747E+06 IC HttLTSIS THOH 27.000 TEUF* 27.000 (A I -2.0008*01 0. 2.000B*01 1.0008*01 6.000E 01 9.9998*03 3 1.2581*0 3.798*01 ****** A*** 1.5818*0 3.798*01 ****** A*** 1.9958*0 3.798*01 ** A 2.5118*01 3.798*01 ****** A + * 3.9818*01 3 .798*01 ****** A+* 5.0118*0 3.798*01 ****** A* + * 6.3098*0 3.798*01 ****** A*** 1.0008*05...

## 136 058

Similar calculations show that the parameters of Q3 are 0.20 pF and Ccs3 0.35 pF. The -3-dB frequency of the circuit can now be estimated by calculating the zero-value time constants for the circuit. First consider C i. The resistance seen across its terminals is given by (7.122), which was derived for the emitter follower. The presence of resistance in series with the collector of Q makes no difference to the calculation because of the infinite impedance of the current generator gm Vi. Thus...

## 14

Also, (W L)21 14 since M26 and M27 are matched. In the example in Section 9.4.3, a compensation capacitor of 3.2 pF provided a 45 phase margin for a feedback factor of unity and a 5-pF load. The DM half-circuits for this example with the independent voltage sources Vsi and Vs2 set to zero are shown in Fig. 12.21 a. Here, we have assumed that CL is much larger than the input capacitance of the CM-sense devices M21 -M2a The two feedback networks connect between the two half-circuits in this...

## 1401118

1.3071-15 sq v hz * 3.615e-08 v rt hz 2.8981+00 -2.1231+00 -2.999e+00 2.907e-03 1.000e+02 3.867e-02 2.586e+03 1.000e+02 2.124e+16 5.605e-13 3.209e-14 0. 0. model id ibs xbd vgs vds vbs vth vdsat beta gam eff 2.931e-03 -8.7621-15 -5.000e-14 1.6881+00 4.1231+00 -8.7621-01 7.0001-01 9.8841-01 6.000e-03 0. 1.241e-15 1.549e-13 1.8501-13 3.7721-14 1.5051-13 1.2411-15 total output boise voltage 4.7861-17 sq v hz 6.9181-09 v rt hz v(5) vi equivali ihput boise at vi freq*7.5ghz equivaleht ihput boise...

## 146

1.3.5 Dependence of Transistor Current Gain pF on Operating Conditions Although most first-order analyses of integrated circuits make the assumption that fiF is constant, this parameter does in fact depend on the operating conditions of the transistor. It was shown in Section 1.3.2, for example, that increasing the value of Vce increases Ic while producing little change in IB, and thus the effective of the transistor increases. In Section 1.3.4 it was shown that as VCe approaches the breakdown...

## 15

Figure 8.21 (c) Small-signal equivalent circuit of the basic amplifier in (b). small capacitors of several picofarads and are included on the chip. They ensure stability of the feedback loop, and their function will be described in Chapter 9. Capacitor Cb is external to the chip and is a large bypass capacitor used to decouple the bias circuitry at the signal frequencies of interest. Bias Calculation. The analysis of the circuit begins with the bias conditions. The bias current levels are set...

## 15 V

Figure 6.59 Circuit for Problem 6.15. Also assume that the biasing is arranged so that Vor 0.2 V for each transistor. Finally, assume that M11 and Mn are biased at the edge of the active region. 6.19 Find the low-frequency voltage gain from variation on each power supply to the op-amp output in Fig. 6.28. Assume that the bias voltages biasi Vbias2, and Vbias3 are produced by the circuit shown in Fig. 6.60, where Mm is the only transistor that operates in the triode region. Assume that the W L...

## 16

This equation shows that the second-harmonic distortion can be reduced by increasing the dc output voltage Vo. This result is reasonable because this distortion stems from the body effect. Therefore, increasing Vo decreases the variation of the source-body voltage compared to its dc value caused by an input with fixed peak amplitude.2 Equation 5.56 also shows that the second-harmonic distortion is approximately proportional to 7, neglecting the effect of y on Vo- Similarly, third-harmonic...

## 169

We will now calculate rc2, assuming a buried-layer sheet resistance of 20 ft Q The distance from the center of the emitter to the center of the collector-contact diffusion is 62 xm, and the width of the buried layer is 41 fxm. The rc2 component is thus, approximately, Here the buried-layer side diffusion was not taken into account because the ohmic resistance of the buried layer is determined entirely by the number of impurity atoms actually diffused see (2.15) into the silicon, which is...

## 16be02

1.59E+02 1.55E*02 1.52E+32-1.IBE*32 1.13E*02 1.38E*C2 1.33E+02 1.27E+C2 1.21E.02 1.1IE*02 1.081*02 1.011*02 9.151*01-8.761*01 8.071*01 7.361*01 6.65E*01 5.931*01 5.201*01 1.IBE. 3.76E.01 3.05E*01 2.35E.01-' mos amp, ezmche skill sigkal bahdwdth as oc vis varies iq1120 m0s2 w 4u l 1d xi 2 3 0 0 mos n 100d l 10 cload 2 0 iooff m 4 3 ik vi 4 0 0.5v ac mdekl mos bbs kp*60u vto 0.7 lambda-0 lb> 0 gamcx-0.4 t tqx'20m coso 300pf cgdoooopp cbfesopt cbs 0k m0e8l m0s2 mos kp*60u vto 0.7 lambda 0 ld 0...

## 17

Op. cit., p. 148. 22. R. S. Muller and T. I. Kamins. Op. cit., pp. 490 196. 23. Y. P. Tsividis. Op. cit., pp. 150-151 and 198-200. 24. R. S. Muller and T. I. Kamins. Op. cit., p. 496. 25. Y. P. Tsividis. Op. cit., p. 151. 26. Y. P. Tsividis. Op. cit., pp. 310-328. 27. R. S. Muller and T. I. Kamins. Op. cit., p. 480. 28. R. S. Muller and T. I. Kamins. Op. cit., p. 482. 29. Y. P. Tsividis. Op. cit., p. 181. 30. Y. P. Tsividis. Op. cit., p. 294. 31. Y. P. Tsividis. Op. cit., p....

## 18

Since the difference in the base-emitter voltages is proportional to the thermal voltage, comparing (4.266) with V0s 0 to (4.249) shows that the gain M here is proportional to (1 + R2 Ri). Rearranging (4.266) gives Vqut VeB2 + ( 1 + )(AV fi) + V05(out) Equations 4.267 and 4.268 show that the output contains an offset voltage that is a factor of (1 + R2fR'i) times bigger than the input-referred offset voltage. Therefore, the same gain that is applied to the difference in the base-emitter...

## 194114

1.1001-04 1.6001-04 1.8001-04 2.0001-04 2.2001-04 2.4001-04 2.600S-04 2.8008-04 3.00CE-04 1.2001-04 3.400E-04 3.6001-04 3.8001-04 4.000E-04 4.200E-04 4.400E-04 4.600E-04 4.8001-04 5-O00E-04 5.2001-04 5.400E-04 5.600E-04 5.8001-04 6.0001-34 6.2031-04 6.4001-04 6.6001-04 6.8031-04 7.0001-04 7.2001-04 7.4001-04 7.6001-04 7.8001-04 8.0001-04 8.2001-04 8.4001-04 8.6001-04 8.8001-04 . 0001-04 9.200E-04 9.400E-04 9.600E-04 9.800E-04 l.OOOE-Ol 031*00-.031*00 031*00 031*00 021*00 .021*00 .021*00 .021*00...

## 1muysis

MVR 2.5171+02 2.5171+02 2.5171+02 2.5171+02 2.S17Z+02 2.5171+02 2.5171+02 2.5171+02 2.5171+02 2.5171+02 2.5171+02 2.5171+02 2.5171+02 2.5172+02 2.517Z+02 3.5171+02 2.5171+02 2.5171+02 2.5171+02 2.5171+02 2.5171+02 2.5171+02 2.5161+02 2.5151+02 2.5111+02 2.5021+03 2.4791+02 2.4241+02 2.2971+02 2.0291+02 1.5691+02 1.0001+02 5.2351+01 2.3801+01 1.0031+01 4.0851+00 1. < 361+00 6.5031-01 RRVI -4.0111-01 -4.0121-01 -4.0141-01 -4.0171-01 -4.0221-01 -4.0291-01 -4.0401-01 -4.0581-01 -4.0871-01...

## 1ocoec2 5000e03 Socoe03 7000803 60ooeo3 5000e03 1000103 3cooe03 2 Oooe03 1000e03

1.OOOE-03 2.OC0E-03 3.OOOE-03 1.C0OE-O3 5.OOOE-03 .OOOE-03 1.0005-33 9.00 jE-03 .003E-33 1,000 s-32 5.508-01-' 5.ISE-01 i 5.138-01 ' 5.3U-01 -5.318-01 5.288-01 ' -5.218-01 < S.UI-01 ' 1.568-01 I -58E-01 1.728+00-' 2 288+00 2.308+00 2.318+30 < 2.32E+00 2.33E+00 ' 2.31E+00 + 2 -34E+00 2.318+00 i 35E+C0- OPERATING POINT INFORMATION TN0M NODE VOLTAGE NOES 'VOLTAGE 0 2 1.779E+00 0 3 1.OOOE-03 0 4 0 5 0. 0 6 1.727E+00 0 9 0 9 2.396E+00 0 100 2.500E+00 0 200 BIPOLAR JUSCTIOH TRANSISTORS SUBCKT...

## 2

Where (1.37) has been substituted for npo. Equation 1.48 shows that 3 . is maximized by minimizing the base width Wb and maximizing the ratio of emitter to base doping densities ND NA. Typical values of fir for npn transistors in integrated circuits are 50 to 500, whereas lateral pnp transistors (to be described in Chapter 2) have values 10 to 100. Finally, the emitter current is -(Ic + IB) - c + n) - F aF The value of aF can be expressed in terms of device parameters by substituting (1.48) in...

## 2 1

A typical value for Kf is 3 X 10 24 V2-F, or 3 x 10 12V2-pF. The equivalent input noise-current generator if for the MOSFET can be calculated by open-circuiting the input of each circuit in Fig. 11.26 and equating the output noise. This gives Since ig and id represent independent generators, (11.70) can be written as

## 2 8 2

for n-chab, 1 - ot*cox 550*1.38e-7 127 ba v**2 for f-chaff, KP dp'cox 250*1.3 e-7 58 oa v**2 . del mos mos level-1 lambda-o.105263 vto-0.7 kp-127u ld-0.12 model pmos pmos level-1 lambda-0.0625 vto--0.7 kp> 58t3 ld-0.18d vidc 3 5 0m vi 5 0 -opticos ncmod nopage .WHJTH OCT-80

## 2 C

Taking the positive square root in the right-most formula in (9.70) yields a value that is larger than one. Adding this value to 1 gives a positive value for the term in parentheses subtracting this value from 1 gives a negative quantity with a smaller magnitude than the sum. Therefore, one zero is in the LHP and has a magnitude greater than gmi (2Cm2). The other zero is in the RHP and has a smaller magnitude than the LHP zero. As a result, the effect of the RHP zero is felt at a lower...

## 2 Cr

One disadvantage of the above method of compensation is that the value of C required is quite large (typically > 1000 pF) and cannot be realized on a monolithic chip. Many general-purpose op amps have unity-gain compensation included on the monolithic chip and require no further compensation from the user. (The sacrifice in bandwidth caused by this technique when using gain other than unity was described earlier.) In order to realize an internally compensated monolithic op amp, compensation...

## 2 L

Where x (VGS - V,) ( CL) as defined for (1.221). If x 1, (1 - x) 1 (1 + x), and ID - - V< > 2 (1226> Equation 1.226 is valid without velocity saturation and at its onset, where (VGS - V,) < s CL. The effect of velocity saturation on the current in the active region predicted by (1.226) can be modeled with the addition of a resistance in series with the source of an ideal square-law device, as shown in Fig. 1.41. Let V'GS be the gate-source voltage of the ideal square-law transistor....

## 20

And thus the collector-base resistance is r l0f30ro 10 X 100 X 20 kO 20 Mil The equivalent circuit with these parameter values is shown in Fig. 1.21. 1.4.8 Specification of Transistor Frequency Response The high-frequency gain of the transistor is controlled by the capacitive elements in the equivalent circuit of Fig. 1.20. The frequency capability of the transistor is most often specified in practice by determining the frequency where the magnitude of the short-circuit, common-emitter current...

## 20 40 6080100

Figure 2.29 Capacitance and depletion-layer width of an abrupt pn junction as a function of applied voltage and doping concentration on the lightly doped side of the junction11 where NB is the doping density in the epi material and VR is the reverse bias on the junction. The nomograph of Fig. 2.29 can also be used to determine the junction depletion-region width as a function of applied voltage, since this width is inversely proportional to the capacitance. The width in microns is given on the...

## 200

* SFETS SUBCKT ELEMENT MODEL IN FIG. 5.31, TBE BIAS VOLTAGE COULD BE ADJUSTED BY TRIAL AND ERROR TO SET THE DRAIN CURRENT OF M3 EQUAL TO 10 MICROAXPS, BUT THIS PROCESS HAT REQUIRE MANT ITERATIONS. SO INSTEAD, K7 AND IB IAS ARE ADDED TO FORM A CURRENT MIRROR TO SET DP THE DC DRAIN M7 2 2 100 100 PMOS W 50U L 1U IBIAS 2 200 9.9U TOE DC OUTPUT IS APPROXIMATELY ZERO. VI 7 200 0.7761 .MODEL PMOS PHOS KP 58U LAHBDA 0.04 GAMfA'0.43 VTO -0.7 LD 0 .MODEL EMOS NWS KP 127U LAMBDA-0.08 GA*1A 0.16 VTO 0.7...

## 2030101

2.991-12-A-2.911-07 > A 1.151-06 t 2 .581-06 1.181-06 . 6.011-06 7.851-06 9.171-06 1.071-05 1.151-05 1.201-05 1.161-05 1.071-05 9.131-06 7.851-06 . 6.001-06 . 1.201-06 2.531-06 1.251-06 1.611-07 A 2.991-12-A-3.2BI-07 J 1.301-06 2.511-06 1.201-06 5.991-06 < 7.851-06 9.121-06 2.098-05 1.151-05 1.201-05 -1.161-05 1.071-05 9.131-06 7.851-06 6.001-06 I.201-06 2.531-06 . 1.251-06 1.611-07 < A 2.991-12-A 3.281-07 A 1.301-06 2.511-06 1.201-06 5.991-06 7.851-06 9.121-06 1.091-05 1.151-05 1.201-05...

## 2076e04 6758e07

1.000E+05 1.2588+05 1.5KE+05 1.9958+05 2.5111+05 3.1621+05 3.9818+05 5.0118+05 6.3091+05 7.9(31+05 1.0001+06 1.2581+06 1.58(1+06 1.9958+06 2.5111+06 3.1621+06 3.9818*06 5.0118*06 6.3098*06 7.9(38*06 1.0008*07 1.2588*07 1.59(8*07 1.9958*07 2.5118-01 3.1528*07 3.9918+07 5.0118+07 i.3091+07 7.9(31+07 1.0001+0 1.25 1+0 1.58(1+0 1.9951+08 2.5111+0 3.1628+08 3.5511+08 5 0112+03 6.3098+0 7.9(38+0 1.0001+09 -1.991+01 - .991*01 -9.001+01 * -9.001*01 -9.CC1+01 -9.001+01 -9.001+01 -9.001+01 -9.001+01 +...

## 21

6x 0W ( 50 )(2 02Xid* J (465 x o'7)'- ciyD Emitter periphery adjacent to a. bait contact is p - 4x 40 > t(m - o Distance from base contact to emitter 5 10 MM ( Series collector resistance F& r each of fwo emitters the effective buried layer dimension* are ' wjgL (W + 2T) 20 > w * + 2(1041*) 5 40 -Mm lBL CL+ 2T) - 4-OM**t 2.(10 Ap*)

## 21 0 13 0 110

Rb 400 br-10 is-6e-18 vat-35 rb-200 bj-20 18-62-18 vaf-30 model pmos1 pnos kp-26u vto--0.7 lambda-0.0625 ld> 0.18u . xl pmos2 fmos kp-26u vto o.7 lambda-0.0244 ld-0.18u lambda1 (dxd dvds) lept 0.04 (1-2*0.18) 0.0625 v*< -1) lambdas s (dxd dvdsj lkft 0.04 (2-2*0.18) > 0.0244 v, -1) the dc input voltage is adjusted bt trial ahd error *to set the dc chttput voltage to zero. the peak input amplitude is set bt trial ahd error 90 that the peak output axplituix is 2 v. vi 9 0 sim -0.8093 2.08...

## 2124105 2000105 9025113 1048105

5.0001-14 9.8071-15 -8.9351-15 4.9931-14 -9.8071-01 -9.8071-01 9.10(1+00 -8.9721-01 -5.0001+00 -9.8071-01 -1.7951-09 -4.9931+00 -7.0001-01 -7.0001-01 7.0001-01 -7.0001-01 1.7951-09 -1.9721-01 s.9811-05 5.3891-04 -8.9351-15 4.1031-14 -5.0001-14 5.0001-14 9.0391-01 -9.8071-01 8.9351-01 -9.8071-01 8.9351-01 -4.1021+00 5.0001+00 -5.0001+00 7.0001-01 -7.0001-01 7.0001-01 -7.0001-01

## 218

Can operate in active or triodc region - M, always operates in tWode region ASSnrnc is in mode region, 2- V < - U I x vp5f 2(V -Vfr) t j Let X . .vo and VD5) must be < Or M, would TVi n, cb The cost of a 40,000 mil* chip is N YWj 47 is obtained from fig p-70) ( i) line cos-t of -t-wo 20,000 mil1 chips,

## 23 High Voltage Bipolar Integrated Circuit Fabrication

Integrated-circuit fabrication techniques have changed dramatically since the invention of the basic planar process. This change has been driven by developments in photolithography, processing techniques, and also the trend to reduce power-supply voltages in many systems. Developments in photolithography have reduced the minimum feature size attainable from tens of microns to the submicron level. The precise control allowed by ion implantation has resulted in this technique becoming the...

## 25

Circuit for Problem 12.23. V0i t Vod(t and Voc(t)l What are Vn(t Vi2 t Vid(t and Vlc(t)7 12.22 The op amp in Problem 12.4 is used with the CMFB scheme shown in Fig. 12.17. The circuit is perfectly balanced except that the CM-sense resistors are mismatched with the upper resistor Rcs 10.1 kil and the lower resistor Rcs2 9.9 kfL Assume the source followers and the CM-sense amplifier are ideal with gains of unity. (a) Compute the gains acms and adm-cms in (12.106). (b) Compute the...

## 2596211 5374e14 6601211 4146e14 1601211 6102e15

1.0002*07 1.1221*07 1.258E+07 1.4122*07 1.584E+07 1.7788*07 1. 958*07 2.2388*07 2.5118*07 2.8188*07 3.1628*07 3.5488*07 3.9818*07 4.4 6E 07 5.0118*07 5.6238*07 6.3098*07 7.079B*07 7.9438*07 8.9122+07 9.9998*07 1.1228*0 1.25B8*0I 1.4128*0 1.5848*0 1.7788*08 1.9958*08 2.2388*08 2.5118*08 2.8188*0 3.1628*0 3.5488*08 3.9811*08 4.4668*08 5.0111*08 5.6238*08 6.3098*08 7.0798*08 7.9438*08 8.9128*0 9.9998*0 1.1228*09 1.2588*09 1.4128*09 1.5848*09 1.7718*09 1.9958+09 2.2388*09 -7.928+01 -7.828*01...

## 26

The gain magnitude at low frequency is thus 36.1 dB and the gain versus frequency on log scales is plotted in Fig. 7.8 for frequencies below and slightly above pi . 7.2.1.2 The MOS Differential Amplifier Differential-Mode Gain A MOS differential amplifier with resistive loads is shown in Fig. 7.9. The differentialmode (DM) ac half-circuit and the corresponding small-signal circuit are shown in Figs. 7.10a and 7.10b, respectively. For compactness, the factor of 1 2 has been omitted from the...

## 2600807 2700807 2109807 2900807 3000807 3100807 3200807 3300807

3.(008-07 3.5008-07 3.6008-07 3.700E-07 3.B00E-07 3.900E-07 4.0008-07 4.100E-07 (.200E-07 4.300E-07 I.1001-07 1.5008-07 I.6008-07 I 7008-07 I 100E-07 1.9008-07 -2.39E-03 -1.228-03 9.(68-03 2.828-02 5.3(8-02 8.0B8-02 1.068-01 1.2(8-01 1.338-01 1.3(8-01 1.278-01-1.158-01 1.028-01 8.981-02 8.118-02 7.708-02 7.7(8-02 8.168-02 8.818-02 9.528-02 1.018-01-1.068-01 1.088-01 1.078-01 1.058-01 1.018-01 9.7(8-02 9.128-02 9.218-02 9.138-02 9.178-02-9.318-02 9.50E-02 9.708-02 9.868-02 9.978-02 1.00E-01...

## 27

.9991+27 .9991+27 .9991+27 .9991+27 .9991+27 .9991+27 .9991+27 ,9991+27 9991+27 9991+27 .9991+27 .9991+27 9991+27 9991+27 .9991+27 9991+27 9991+27 ,9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 9991+27 000 pi 27.000 mii -1.5921+25 -1.5921+25 -1.5921+25 -1.5921+25 -1.5921+25 -1.5921+25 -1.5921+25 -1.5921+25 -1.5921+25...

## 27000

2.21J+00 * 2.181*00 *. . * 2.181*00 * * * * * * i * 2.121*00 * * * * . 2.2001*00 2.2101*00 2.2201*00 2.2301*00 2.2101*00 * OPERATING POIHT nffORMATIOtf THMI 27 7BMP 27 m VOLTAOE BOSS VOLTAGE SOBS VOLTAGE +0 1 * 2.2681+00 0 2 * 2.0001*00 0 4 7.3S0E-01 +0 100 > 3.000E+00

## 27000 Temp 27000

1.001-00-j.HE-01 9-531-01 9.531-01 9.711-01 1.001*00 1.031-00 l.tHl'K 1.041-00 1-031-00 1.00E-00-9.71E-01 9.531-01 9.531-01 9.71E-01 1.00E-00 1.03E+00 1.0IE-00 1.01E-00 1.03E-00 1.001-00- FOORIER COMPOHHITS OP TRAHSIBR RESPONSE V(2) DC CCMPUBEHT - l.OOODtOO HARMCHIC FREQUENCY FOORIER NORMALIZED PHASE NORMALIZED NO (HZ) CCMPOHEOT COMPONENT (DEG) PHASE (DEG) FOORIER COMPOHHITS OP TRAHSIBR RESPONSE V(2) DC CCMPUBEHT - l.OOODtOO HARMCHIC FREQUENCY FOORIER NORMALIZED PHASE NORMALIZED NO (HZ)...

## 282

From (7.74b) and (7.74c), the coefficients of the denominator of the transfer function are 2k (7.33p + 0.5p) + 190 (7.33p) + 190 (0.0282)(2k)(0.15p) a - -----------s 0.324 ns 190(2k) (0.5p)(7.33p) + (7.33p)(0.15p) 1 + (0.0282)(2k) Using the quadratic formula to solve for the poles, we find that the poles are pii2 -5.1 x 109 2.3 x 109 rad s The poles and zero are fairly close together, as shown in Fig. 7.19a. The gain magnitude and phase are plotted in Fig. 7.19. The -3-dB bandwidth is 1.6 GHz....

## 2x

MODEL NPH NPN RB*200 BF-200 VAF 130 IS 5E-15 MODEL PHP PHP RB 300 BF 50 VAF 50 IS 2E-15 MODEL NPH NPN RB*200 BF-200 VAF 130 IS 5E-15 MODEL PHP PHP RB 300 BF 50 VAF 50 IS 2E-15 A DC IMPOT VOLTAGE OF ABOUT 1 MV IS REQUIRED TO FORCE ALL THE TRANSISTORS TO OPERATE IN THE FORWARD-ACTIVE REGION. VIDC 3 5 1M VI 5 0 . OPTIOKS NQMOD NOPAGE .WIDTH & 7TX80 .OP

## 2x4x5

These values are marked on the root locus of Fig. 9.39. In this example, it is useful to compare the prediction of instability at Tq 13.2 with the results using the Nyquist criterion. The loop gain in the frequency domain is A series of trial substitutions shows that LT(jco) -180 for co 3.8 X 106 rad s. Note that this is close to the value of 4 X 106 rad s where the root locus was assumed to cross the jco axis. Substitution of a) 3.8X106 in (9. Ill) gives, for the loop gain at that frequency,

## 3

1.000E+05 1.2562+05 1.5B4E 05 1.9955*05 2.5115*05 3.1625+05 3-981E+05 5.0I1E-05 6.309E+05 7.9432+05 1.0005+06 1.253E+06 1.5812*05 1.9955+06 2.5115.06 3.162E+06 3.9B15-06 5.0115*06 6.309E*06 7 .9435+05 1.0005*07 1.2535+0 i.5845+07 1.9955+07 2.5115*0'' 3.1S25-C7 3.9815*07 5.0115*07 6.3095*0 7 9435*0 I. SBE+08 1.534E*08 1.9955*09 2.5115-09 3.1625+03 3.9915+09 5.0115+08 6.3095+08 7.5435*08 1.0005*09 1.775*02 .775+02 1.765*02 1.755*02 .745*02 1-73E.02 1.71E.02 1.635*02 1.575*02 1.635*02 1.595*02-1....

## 3 V

Figure 8.50 Balanced series-shunt feedback amplifier. 8.14 Replace npn transistors Q -Qi in Fig. 8.49 with NMOS transistors My-Mj, and replace the pnp transistor Qi with PMOS transistor Af< . Also, replace the 1.25 kfl resistor in the drain of M with a 4.35 kfl resistor. Repeat the calculations and simulations in Problem 8.13. For all transistors, use W L 100, y 0, and A 0. Also, V,n -vtp 1 V, k'n 60 (xAV2, and k'p 20 xAA 2. 8.15 A balanced monolithic series-shunt feedback amplifier is shown...

## 306801

-7.478-02-1.101-02 1.578-01 2.72E-01 3.881-01 5.04E-01 5.211-01 1.051-0 2.B91-01 1.711-01 5.BDE-02 -5.77E-02 -1.731-01 -2.891-01 -1.051-01 -5.201-01 -5.051-01 -3.891-01 -2.711-01 -1.581-01 -1.21E-02-7.338-02 1.891-01 3.051-01 1.201-01 5.368-01 1.908-01 3.718-01 2.598-01 1.431-01 2.731 -1.848-02 -2.041-01 -3.201-01 -4.351-01 < -5.511-01 -4.711-01 -3.551-01 . -2.401-01 < -1.241-01 -9 . 1.071-01 * 2.231-01 3.39E-01 1.541-01 5.7CE-01 4.551-01 3.108-01 2.218-01 1.08E-01 -7.50E-03 -1.23E-C1...

## 31

Since Vds3 Vd32, Vm y,32 Vtn, and (W L)3l (W L)32. This equation shows that Icm is dependent on the CM output voltage and independent of the differential output voltage because changes in the drain currents in M31 and M32 due to nonzero V* are equal in magnitude and opposite in sign. Therefore, these changes cancel when the drain currents are summed in (12.62). Applying KVL around the lower transistors A 30-M35 gives Vds3l Vds35 + Vgs33 - (12.63) Assuming that I0 33, we have Vgs30 V .33, and...

## 3158c 1398c

1.0008-08 -5.578-02 1.5008-08 -2.508-01 2.0008-08 -1.158-01 * 2.5008-08 -6.108-01 3.0008-08 -S.318-01 3.5008-08 -1.02B+00 1.0008-08 -1.228+00 * 1.5008-08 -1.398*00 * 5.0008-08 -1.15E*00-+A-5.5008-08 -1.178*00 A 6.0008-08 -1.178*00 +A 6.5008-08 -1.188+00 A 7.0008-08 -1.188+00 A 7.5008-08 -1.188*00 A 8.0008-08 -1.198*00 A 8.5008-08 -1.198+00 A 9.0008-08 -1.198+00 A 9.5008-08 -1.198+00 > 1.0008-07 -1.19E+00-A-- small-sigkal transfer characteristics -4-733e+04 9.999e+19 6.296e+04 5at k, (Vq s,-Hj...

## 3160

This can be referred back to the input of the complete circuit of Fig. 11.38a in the standard manner. Consider the contribution to the equivalent input noise voltage. A voltage applied at the input of the full circuit of Fig. 11.38a gives Equating output noise current in (11.112) and (11.113), we obtain an equivalent input noise voltage due to i2oA as Thus the active load causes a contribution of 9.64 kil to the equivalent input noise resistance of the 741 op amp. The remaining important...

## 3300804

3 4C38-34 3.5008-04 3.6008-04 3.7308-04 3.1308-04 3.9008-04 4.0008-04 4.1008-34 4.2308-04 4.3008-04 4.4008-04 4.5002-34 4.60C8-04 4 .008-34 4.5008-04 4.9338-04 9.968-01 3.558-06 9.358-06 1.618-05 2.558-05 3.538-05 4.618-05 5.798-05 7.048-05 8.378-05 9.778-05-1.128-04 1.288-04 1.438-04 1.608-04 1.nl8-04 1.948-04 2.128-04 2.308-04 2.498-04 2.688-04-2.878-04 3.068-04 3.268-04 3.478-04 3.578-04 3.888-04 4.098-04 4.318-34 4.528-04 4.748-04 4.968-04 5.188-04 5.438-04 5.638-04 5.368-04 6.098-04...

## 35

This last expression shows that the op-amp tail current Icms consists of a constant term 2 i plus a term that depends on Voc-yCM-If 1 31 ld41 h, then KCL requires that Icms 2 i. Using this value in (12.66) gives Voc Vcm, as desired. In practice, mismatches can cause Voc to deviate from Vcm For example, if the drain currents in M3 and M4 are larger than 11, then (12.66) shows that Voc must be larger than VCm to force lcms to be larger than 2 Ix. To see that the CMFB loop here is a negative...

## 35 30 25 20 15 10 5

Figure 2.63 (a) Plan view and cross section of polysilicon resistor. resistor described in Section 2.6.2 and shown in Fig. 2.42. It displays large tolerance, high voltage coefficient, and high temperature coefficient relative to other types of resistors. Higher sheet resistance can be achieved by the addition of the pinching diffusion just as in the bipolar technology case. MOS Devices as Resistors. The MOS transistor biased in the triode region can be used in many circuits to perform the...

## 3500801

3.5508-01 -3.668-01 3.6008-01 -1.588-01 > 3.6508-01 1.B9B-02 3.7008-01 2.568-01 3.7508-01 1.638-01 3.8008-01 1.798-01 3.8508-01 2.728-01 3.9008-01 6.118-02 * 3.9508-01 -1.438-01 ' 4.0008-04 -3.508-01 4.0508-01 -5.578-01 4.1008-01 -3.858-01 4.1508-04 -1.788-01 1.2008-01 2.918-02 -1.2508-01 2.378-01 1.3008-01 1.118-01 1.3508-01 4 .998-01 4 4008-04 2.918-01 4.4538-04 8.118-02 -1.5008-01 -1 * 1.5508-04 -3.308-01 -4.6008-01 -5.378-01 1.6508-01 -1.058-31 1.7008-01 -1.988-01 1.7508-01 9.588-03...