102 Operating Principles

FET 1 and FET 2 (the power switches) turn on, or off, simultaneously. When the devices are switched on, the primary supply voltage Vci will be applied across the transformer primary, and the starts of all windings will go positive. Under steady-state conditions, a current will have been established in the output choke LI by previous cycles, and this current will be circulating by flywheel action in the choke LI, capacitor CI, and load, returning via the flywheel diode When the secondary emf is...

110 Specifica Tion Notes

The designer should be alert to the tendency for specifications to escalate. When a flyback converter is to be considered and potential requirements are large, costs are often particularly sensitive. The designer should establish with the customer the real limitations of the application. It may well be that a typical performance of 6 regulation on the auxiliary outputs of a multiple-output unit would be acceptable. This allows a semiregulated flyback system to be used. To guarantee a result of...

1101

EES. 1.12.1 Characteristics of a (ypical undervoltage transient protection circuit. (a) Load current transient. (b) Typical undervoltage transient excursion without protection circuit fitted, (c) Undervoltage transient excursion with protection circuit fitted. EES. 1.12.1 Characteristics of a (ypical undervoltage transient protection circuit. (a) Load current transient. (b) Typical undervoltage transient excursion without protection circuit fitted, (c) Undervoltage transient excursion with...

114 Crowbar Performance

More precise crowbar protection circuits are shown in Fig. 1.11.1b and c. The type of circuit selected depends on the performance required. In the simple crowbar, there is always a compromise choice to be made between ideal fast protection (with its tendency toward nuisance operation) and delayed operation (with its potential for voltage overshoot during the delay period). For optimum protection, a fast-acting, nondelayed overvoltage crowbar is required. This should have an actuation voltage...

115 Limitations Of Simple Crowbar Circuits

The well-known simple crowbar circuit shown in Fig. 1.11. la is popular for many noncritical applications. Although this circuit has the advantages of low cost and circuit simplicity, it has an illdefined operating voltage, which can cause large operating spreads. It is sensitive to component parameters, such as temperature coefficient and tolerance spreads in the zener diode, and variations in the gate-cathode operating voltage of the SCR. Furthermore, the delay time provided by Cl is also...

1168

When the rectified waveform is compared to a reference voltage, the change in supply voltage shows up as an increase in the time At taken for the rectified voltage to exceed the reference value. This change can be used to indicate a probable failure before the full half cycle has been established. This method gives the earliest possible warning of power brownout or failure. Figure 1.21.66 shows a suitable circuit. This circuit operates as follows. The line input is bridge-rectified by diodes Dl...

118 Selecting Fuses For Scr Crowbar Overvoltage Protection Circuits

In the event of an overvoltage stress condition caused by the failure of the series regulator in a linear power supply, the crowbar SCR will be required to conduct and clear the stress condition by blowing the series protection fuse. Hence, the designer must be confident that the fuse will open and clear the ulty circuit before the SCR is destroyed by the fault current. If a large amount of energy is dissipated in the junction of the SCR within a short period, the resultant heat cannot be...

119 Type 3 Overvoltage Protection By Voltage Limiting Techniques

In switchmode power supplies, the crowbar or clamp voltage protection techniques tend to be somewhat less favored because of their relatively large size and dissipation. By its nature, the off-line switchmode power supply tends to fail safeM that is, to a zero or low-voltage condition. Most failure modes tend to result in zero output voltage. Since the high-frequency transformer provides galvanic isolation between the input supply and the output lines, the need for crowbar-type overvoltage...

121 Introduction

The need for undervoltage protection is often overlooked in system design. In most power systems, a sudden and rapid increase in load current (for example, inrush currents to disk drives) results in a power supply line voltage dip. This is due to the rapid increase in current during the transient demand and the limited response time of the power supply and its connections. Even when the transient performance of the power supply itself is beyond reproach, the voltage at the load can still dip...

121 Output Ripple and Noise

Where very low levels of output ripple are required, the addition of a small LC noise filter near the output terminals will often eliminate the need for expensive low-ESR capacitors in the main secondary reservoir positions. For example, a typical 5-V 10-A supply may use the highest-quality low-ESR capacitors in positions Ci, C2, and C3 of the single-stage filter shown in Fig. 2.1.1, but this will rarely give a ripple figure of less than 100 mV. However, it is relatively easy to keep ripple...

1210 Optimum Flux Density

The choice of optimum flux density 5opt will be a matter for careful consideration. Unlike with the flyback converter, both quadrants of the BIH loop will be used, and the available induction excursion is more than double that of the flyback case. Consequently, core losses are to be considered more carefully as these may exceed the copper losses if the full induction excursion is used. For the most efficient design, the copper and core losses should be approximately equal. Figure 2.12.2 shows...

1212 Calcula Ting Primary Turns

Once optimum core size and peak flux density have been selected, the primary turns may be calculated. The transformer must provide full output voltage at minimumline input. Under these conditions, the power pulse will have its maximum width of 16.5 (jls. Hence the minimum primary turns are calculated for this condition. With 90 V rms input to the voltage doubling network, the DC voltage will be approximately 222 V. (See Part 1, Chap. 6.) Consider one half cycle of operation. The capacitors CI...

1215 Control And Drive Circuits

The control and drive circuits used for this type of converter are legion. They range from fully integrated control circuits, available from a number or manufacturers, to the fully discrete designs favored by many power supply engineers. A discussion of suitable drie circuits will be found in Part 1, Chaps. 15 and 16. For reliable operation, the drive and control circuits must provide the following basic functions 1. Soft start. This reduces inrush current and turn-on stress, and helps to...

1216 Flux Doubling Effect

The difference in the operating mode for'the single-ended transformer and the push-pull balanced transformer is not always fully appreciated. For the single-ended forward or flyback converter, only one quadrant of the BIH loop is used, there is a remnant flux B and the remaining range of induction is often quite small. (Figure 2.9,2a and b shows the effect well.) in the push-pull transformer, it is normally assumed that the full BIH loop may be used and that B will be incremented from Bmto each...

122 Operating Principles

Figure 2.12,1 a shows the general arrangement of power sections for the half-bridge push-pull converter. The switching transistors Ql and Q2 form only one side of the bridge-connected circuit, the remaining half being formed by the two capacitors Cl and C2. The major difference between this and the full bridge is that the primary of the transformer will see only half the supply voltage, and hence the current in the winding and switching transistors will be twice that in the full-bridge case....

122 Undervoltage Suppressor Performance Parameters

Figure 1.12. la, b, and c shows the typical current and voltage waveforms-that may be expected at the load end of the DC output lines from a power supply when a large transient load is applied by the load. Figure 1.12.1a shows a large transient load current demand during the period from t, to t,. Figure 1.12.1& shows the undervoltage transient that might typically be expected at the load during this transient. (Assume that the voltage dip is caused by the resistance and inductance of the...

124 Practical Circuit Description

Figure 1.12.4 shows a practical implementation of this technique. In this circuit, switch SW1 or Q1 is replaced by Darlington-connected transistors Q3 and Q4. These transistors operate as a switch and linear regulator. FIG. 1.12.4 Example of an undervoltageprotection circuit. FIG. 1.12.4 Example of an undervoltageprotection circuit. Although Q3 and Q4 are now shown positioned between the two capacitors CI and C2, it has been demonstrated above that since they still form a series network, their...

124 Problem Areas

The designer must guard against a number of possible problems with this type of converter. A major difficulty is staircase saturation of the transformer core. If the average volt-seconds applied to the primary winding for all positive-going pulses is not exactly equal to that for all negative-going pulses, the transformer flux density will increase with each cycle (staircase) into saturation. The same effect will occur if the secondary diode voltages are unbalanced. As storage times and...

125 Currentmode Control And Subharmonic Ripple

A coupling capacitor Cx is often fitted to prevent a DC path through the transformer winding when the supply is linked for 110-V operation. This capacitor is intended to prevent staircase saturation by blocking DC current in the transformer primary. Unfortunately, it can also introduce an undesirable effect, characterized by alternate cycles being high-voltage, narrow pulse width and low-voltage, wide pulse width as a result of the capacitor Cx developing a DC bias. This unbalanced operation...

126 Crossconduction Prevention

Cross conduction can be a major problem in the half-bridge arrangement. Cross conduction occurs when both Ql and Q2 are on at the same instant, usually as a result of excessive storage time in the off'-going transistor. This fault applies a short circuit to the supply lines, usually with disastrous results. Two methods are suggested to stop this effect. The simple approach is to apply a fixed end stop to the drive pulse width so that the conduction angle can never be wide enough to allow cross...

126 Transient Behavior

When a transient current demand occurs, it will reduce the voltage across the load and input terminals 1 to 6. The negative end of C3 will track this change, taking the emitter of Ql negative. After a few millivolts change, Ql will start to turn on, bringing Q2 into conduction. Q2 will drive the Darlington-connected linear regulator transistors Q3 and Q4 into conduction. This action progressively connects Cl and C2 in series, driving current into the output terminals 1 to 6 to prevent any...

128 Soft Start

When the converter is first switched on, the drive pulses should be progressively increased to allow a slow buildup in output current and voltage. This is known as soft start. If this soft-start action is not provided, there will be a large inrush current on initial switch-on, with an overshoot in output voltage also, the transformer may saturate as a result of flux doubling effects. (See Part 3, Chap. 7.) The soft-start action should always be invoked following a shutdown of the converter for...

13 Operating Modes

Two modes of operation are clearly identifiable in the flyback converter 1. Complete energy transfer (discontinuous mode), in which all the energy that was stored in the transformer during an energy storage period (on period) is transferred to the output during the flyback period (off period). 2. Incomplete energy transfer (continuous mode), in which a part of the en-ergy scored in the transformer at the end of an on period remains in the transfohner at the beginning of the next on period.

131 Introduction

In computer and professional-grade power supplies, it is normal practice to provide full overload protection. This includes short-circuit protection and current limits on all outputs. The protection methods take many forms, but in all cases the prime function is to protect the power supply, irrespective of the value or duration of the overload, even for continuous short-circuit conditions. Ideally the load will also be protected. To this end the current limit values should not exceed the...

1321 General Conditions

Figure 2.13.1 shows the power section of a typical off-line bridge converter. Diagonal pairs of switching devices are operated simultaneously and in alternate sequence. For example, Ql and Q3 would both be on at the same time, followed by Q2 and Q4. In a pulse-width-controlledsystem, there will be a period when all four devices will be off. It should be noted that when Q2 and Q4 are on, the voltage across the primary winding has been reversed from that when Ql and Q3 were on. In this example, a...

133 Type 1 Overpower Limiting

The first type is a power-limiting protection method, often used in flyback units or supplies with a single output. It is primarily a power supply short-circuit protection technique. This and the methods used in types 2 and 4 are electronic, and depend on the power supply remaining in a serviceable condition. The supply may be designed to shut down or self-reset if the overload is removed. In this type of protection, the power (usually in the primary side of the con- verter transformer) is...

134 Type 1 Forma Primary Overpower Limiting

In this form of power limiting, the primary power is constantly monitored. If the load tries to exceed a defined maximum, the input power is limited to prevent any further increase. Usually, the shape of the output current shutdown characteristic is poorly defined when primary power limiting is used on its own. However, because of its low cost, primary power limiting has become generally accepted in lower-power, low-cost units (particularly in multi-output flyback power supplies). It should be...

135 Type 1 Form B Delayed Overpower Shutdown Protection

One of the most effective overload protection methods for low-power, low-cost supplies is the delayed overpower shutdown technique. This operates in such a way that if the load power exceeds a predetermined maximum for a duration beyond a short defined safe period, the power supply will turn off, and an input power off-on cycle will be required to reset it to normal operation. Not only does this technique give the maximum protection to both power supply and load, but it is also the most...

136 Type 1 Form C Pulsebypulse Overpowercurrent Limiting

This is a particularly useful protection technique that will often be used in addition to any secondary current limit protection. * The input current in the primary switching devices is monitored on a real-time basis. If the current exceeds a defined limit, the on pulse is terminated. With discontinuous flyback units, the peak primary current defines the power, and hence this type of protection becomes a true power limit for such units. With the forward converter, the input power is a function...

142 Foldback Principle

Figure 1.14.1 shows a typical reentrant characteristic, as would be developed measured at the output terminals of a foldback-limited power supply. A purely resistive load will develop a straight load line (for example, the 5-fl load line shown in Fig. 1.14.1). A resistive load line has its point of origin at zero, and the current is proportional to voltage. As a resistive load changes, the straight line (which vill start vertically at zero load i.e., infinite resistance) will swing clockwise...

143 Foldback Circuit Principles As Applied To A Linear Supply

FOLDBACK output CURRENT LIMITING 1.115 14. FOLDBACK output CURRENT LIMITING 1.115 REGULATOR TRANSISTOR AND CURRENT LIMIT CIRCUIT REGULATOR TRANSISTOR AND CURRENT LIMIT CIRCUIT ED3. 1.14.2 (a) Foldback current limit circuit, (b) Regulator dissipation with reentrant protection. ED3. 1.14.2 (a) Foldback current limit circuit, (b) Regulator dissipation with reentrant protection. put voltage is zero (output short circuit). At short circuit, the current in R1 is very small, and the voltage across...

144 Lockout In Foldback Currentlimited Suppues

With the resistive load (the straight-line loads depicted in Figs. 1.14.1 and 1.14.3), there can only be one stable point of operation, defined by the intersection of the NON LINEAR LOAD LINE (LOCK OUT AT 'P2') EK3. 1.14.3 Overload and start-up characteristics of a foldback, current-limited supply, showing performance for linear and nonlinear load lines. NON LINEAR LOAD LINE (LOCK OUT AT 'P2') EK3. 1.14.3 Overload and start-up characteristics of a foldback, current-limited supply, showing...

145 Reentrant Lockout With Crossconnected Loads

Lockout problems can occur even with linear resistive loads when two or more foldback-limited power supplies are connected in series. (This series connection is often used to provide a positive and negative output voltage with respect to a common line.) In some cases series power supplies are used to provide higher output voltages. Figure 1.14,5a shows a series arrangement of foldback-limited supplies. Here, positive and negative 12-V outputs are provided. The normal resistive loads R1 and R2...

15 Energy Storage Phase

The energy storage phase is best understood by considering the action of the basic single-output flyback converter shown in Fig. 2.L2. When transistor Ql is turned on, the start of all windings on the transformer FIG. 2.1.2 Simplified power section of a flyback (buck-boost) converter. will go positive. The output rectifier diode D1 will be reverse-biased and will not conduct therefore current will not flow in the secondary while Q1 is conducting. During this energy storage phase only the...

153 Incorrect Turnoff Drive Waveforms

Surprisingly, it is the application of energetic and rapid reverse base drive during the turn-off edge which is the major cause of secondary breakdown failure of high-voltage transistors with inductive loads. Under aggressive negative turn-off drive conditions, carriers are rapidly removed from the area immediately adjacent to the base connections, reverse-biasing the base-emitter junction in this area. This effectively disconnects the emitter from the remainder of the chip. The relatively...

155 Correct Turnon Waveform

During the turn-on edge, the reverse of the above turn-off action occurs. It is necessary to get as much of the high-resistance region of the collector conducting as quickly as possible. To achieve this, the base current should be large, with a fast-rising edge thus earners are injected into the high-resistance region of the collector as quickly as possible. The turn-on current at the beginning of the on period should be consider ably highe than that necessary to maintain saturation during the...

156 Antisaturation Drive Techniques

FIG. 1.15.1 (a) Base drive current shaping for high-voltage bipolar transistors, (b) Collector voltage, collector current, base drive current, p-a base emitter voltage waveforms. FIG. 1.15.1 (a) Base drive current shaping for high-voltage bipolar transistors, (b) Collector voltage, collector current, base drive current, p-a base emitter voltage waveforms. The base drive current waveforms are shown in Fig. 1.15, b. Although it is not essential to profile the drive current waveform for all types...

157 Optimum Drive Circuit For Highvoltage Transistors

A fully profiled base drive circuit is shown in Fig. 1.15. la, and the associated drive waveforms are shown in Fig. 1.15.16. This drive circuit operate as follows. When the drive input to point A goes positive, current will initially flow via CI and D1 into the base-emitter junction of the switching transistor Ql. The initial current is large, limited only by the source resistance and input resistance to Ql, and Ql will turn on rapidly. As CI charges, the voltage across Rl, R2, C2, and Lb will...

16 Energy Transfer Modes Flyback Phase

When Q1 turns off, the primary current must drop to zero. The transformer ampere-turns cannot change without a corresponding change in the flux density LB. As the change in the flux density is now negative-going, the voltages will reverse on all windings (flyback action). The secondary rectifier diode Dl will conduct, and the magnetizing current will now transfer to the secondary. It will continue to flow from start to finish in the secondary winding. Hence, the set-ondary (flyback) current...

164 Turnoff Action

When Q2 is turned on again, at the end of a conducting period of Ql, the voltage on all windings is taken to near zero by the clamping action of Q2 and D1 across the clamp winding S2. The previous proportional drive current from PI is now transformed into the loop S2, Dl, and Q2, together with any reverse recovery current from the base-emitter junction of Ql via SI (less the current transformed from P2 as a result of conduction in Rl). Hence the base drive is removed, and Ql turns off. As...

165 Drive Transformer Restoration

For the first part of the on period of the driver transistor Q2, Dl and S2 will be conducting. However, when Ql has turned off and the recovery current in the base-emitter junction of Ql has fallen to zero, S2 and hence Dl will become reversed-biased as a result of the voltage applied to winding P2 via Rl. The start of all windings will now go negative, and current will build up in winding P2, resetting the core back toward negative saturation. At saturation, the current in P2 and Q2 is limited...

166 Widerange Proportional Drive Circuits

Where the range of input voltage and load are very wide, the circuit shown in Fig. 1.16.1 will have some limitations, as follows. When the input voltage is low, the duty-cycle will be large, and Ql may be on for periods considerably exceeding 50 of the total period. Further, if the minimum load is small, LI will be large to maintain continuous conduction in the output filter. Under these conditions, the collector current is small, but the on period is long. During the long on period, a...

168 Turnon Action

When Q2 is turned off, the starts of all windings will go positive by flyback action, and Q1 will be turned on. Regenerative drive from PI and Si maintains the drive, holding Q1 and Q3 on apd rapidly recharging Cl. This action is maintained until Q2 is turned on again to complete the cycle. The advantage of this arrangement is that the core can be reset rapidly by using a high auxiliary supply voltage without excessive dissipation in R1 and Q2. Hence, in this circuit the conflict between...

169 Proportional Drive With Highvoltage Transistors

IfQl is a high-voltage transistor, it is probable that some shaping of the base drive current will be required for reliable and efficient operation, as shown in Sec, 15.1 of Part 1. Figure 1.16.3 shows a suitable modification to the drive circuit in Figure 1.16.2 for high-voltage transistors base drive shaping has been provided by R4, D3, C2, R3, and Lb. EDS. 1.16.3 Push-pull-type proportional drive circuit with special drive current shap ing for high-voltage transistors. EDS. 1.16.3...

171 Complete Energy Transfer

If the flyback current reaches zero before the next on period of Ql, as shown in Fig. 2.1.5a, the system is operating in a complete energy transfer mode. That EE3. 2.1.5 (a) Primary current waveform I and secondary current waveforms I, (discontinuous-mode) operation, ib) Primary and secondary waveforms for incomplete energy transfer (continuous-mode) operation. EE3. 2.1.5 (a) Primary current waveform I and secondary current waveforms I, (discontinuous-mode) operation, ib) Primary and secondary...

172 I ncom plete Energy Transfer

If, in the circuit example shown in Fig. 2.1.2, the on period is increased and the off' period correspondingly decreased, more energy is stored in he transformer during the on period. For steady-state operation, this extra energy must be extracted during the off' period. If the input and output voltages are to be maintained constant, it will be shown that the load current must be increased to remove the extra energy. The slope of the input and output current characteristics cannot change,...

18 snubber Networks 186 Turnoff Dissipation In Transistor Q1

By the same logic as used above (although the waveform is inverted), CI and transistor Q1 both see the same mean current and voltage during the turn-off period. Hence, the dissipation in the transistor during the turn-off period t, to t, will be the same as the energy stored in Cl at the end of the turn-off period (f2). power dissipated in Q1 during the off period, mW snubber capacitance, (JiF Vcco rating of transistor (70 Vceo is the chosen maximum voltage at lc 0) frequency, kHz

18 Transfer Function Anomaly

Ve effective volume of core and air gap 1. MULTIPLE OUTPUTFLYBACK SWITCHMODE POWER SUPPLIES 2.13 This power is proportional to the shaded area to the left of the BIH curve in Fig. 2.1.6 it is clearly larger for the example in Fig. 2.1.66 (the incomplete energy transfer case). Much of the extra energy is stored in the air gap consequently, the size of the air gap will have a considerable effect upon the transmissible power. Because of the very high reluctance of the air gap, it is quite usual to...

181 Introduction

Snubber networks (usually dissipative resistor-capacitor diode networks) are often fitted across high-voltage switching devices and rectifier diodes to reduce switching stress and EMI problems during turn-off or turn-on of the switching device. When bipolar transistors are used, the snubber circuit is also required to give load line shaping and ensure that secondary breakdown, reverse bias, safe operating area limits are not exceeded. In off-line flyback converters, this is particularly...

1810 The Wea Ving Lowloss Snubber Diode

As shown above, to reduce secondary breakdown stress during the turn-off of high-voltage bipolar transistors, it is normal practice to use a snubber network. Unfortunately, in normal snubber circuits, a compromise choice must be made between a high-resistance snubber (to ensure a low turn-on current) and a low-resistance snubber (to prevent a race condition at light loads where narrow pulse widths require a low CR time constant). This paradox often results in a barely satisfactory compromise....

182 Snubber Circuit With Load Line Shaping

Figure 1.18. la shows the primary of a conventional single-ended flyback converter circuit PI, Q1 with a leakage inductance energy recovery winding and diode P2, D3. Snubber components Dl, Cl, and R1 are fitted from the collector to the emitter of Ql. Figure 1.18,16 shows the voltage and current waveforms to be expected in this circuit. If load line shaping is required, then the main function of the snubber components is to provide an alternative path for the inductively maintained primary...

183 Operating Principles

During the turn-off edge of Ql, under steady-state conditions, the action of the circuit is as follows. As Q1 starts to turn off at tl (Fig. 1.18.1A), the primary and leakage inductance of Tl will maintain a constant primary current lP in the transformer primary winding. This will cause the voltage on the collector of Ql to rise (tl to t2), and the primary current will be partly diverted into Dl and CI (ls) (Cl being discharged at this time). Hence, as the current in Ql falls, the inductance...

184 Establishing Snubber Component Values By Empirical Methods

Referring again to Fig. 1.18.1a,, unless the turn-off time of Ql is known (for the maximum collector current conditions and selected drive circuit configuration), the optimum choice for Cl will be an empirical one, based upon actual measurements of collector turn-off voltages, currents, and time. The minimum value of Cl should be such as to provide a safe voltage margin between the Vceo rating of the transistor and the actual measured collector voltage at the instant the collector current...

185 Establishing Snubber Component Values By Calculation

Figure 1.18.1b shows typical turn-off waveforms when the snubber network Dl, CI, R1 shown in Fig. 1.18.1 is fitted. In this example, CI was chosen such that the voltage on the collector Vce will be 70 of the VceQ rating of Ql when the collector current has dropped to zero at time f2. Assuming that the primary inductance maintains the primary current constant during the turn-off edge, and assuming a linear decay of collector current in Ql from t, to t2, the snubber current I, will increase...

188 Dissipation In Snubber Resistor

The energy dissipated in the snubber resistor during each cycle is the same as the energy stored in Cl at the end of the off' period. However, the voltage across Cl depends on the type of converter circuit. With complete energy transfer, the voltage on Cl will be the supply voltage Vcc, as all flyback voltages will have fallen to zero before the next on period. With continuous-mode operation, the voltage will be the supply voltage plus the reflected secondary voltage. Having established the...

189 Miller Current Effects

When measuring the turn-off current, the designer should consider the inevitable Miller current that will flow into the collector capacitance during the turn-off edge. This effect is often neglected in discussions of high-voltage transistor ac tion. It results in an apparent collector-current conduction, even when Ql is fully turned off. Its magnitude depends on the rate of change of collector voltage (dVcldt) and collector-to-base depletion capacitance. Further, if the switching transistor Ql...

19 Transformer Throughput Capability

It is sometimes assumed that a transformer operating in the complete energy transfer mode has greater transmissible power than the same transformer operating in an incomplete transfer mode. (It sounds as if it should.) However, this is true only if the core gap remains unchanged. Figure 2,1,6a and b shows how, by using a larger air gap, the same transformer may be made to transfer more power in the incomplete transfer mode than it did previously in the complete transfer mode (even with a...

193 Crosscoupled Inhibit

Figure 1.19.3 shows the basic elements of a dynamic cross-coupled, cross-conduction inhibit technique, applied in this example to a push-pull converter. In a similar way to the previous example, if, in the push-pull converter, transistors Ql and Q2 are turned on at the same instant, the primary winding of the transformer T2 will be short-circuited and very large collector currents will flow in the transistors, probably with catastrophic results. In Figure 1.19.3, cross conduction is prevented...

1X04 577mm

This is a relatively large core, and for economy in low-current applications more primary turns may be used. For example, 5 turns on the primary and a core of V of the previous area will give the same delay. The area would now be A, 11.4 mm2, and a TDK T7-14-3.5 or similar toroid would be suitable. It may be necessary to fit a resistor (Rl, R2) across the rectifier diodes D1 and D2 to allow full restoration of the core during the off' period, as the leakage current and recovered charge from D1...

2010 Main Output Inductor Values Buck Regulators

In general, the main inductance LI in the output of a buck regulator filter circuit should be as small as possible to give the best transient response and minimum cost. If a large inductance is used, then the power supply cannot respond rapidly to change5 in load current. At the other extreme, too low an inductance Vflll result in very large ripple currents in the output components and converter circuits which will degrade the efficiency. Further, discontinuous operation will occur at light...

2011 Design Example

Assume that a design is required for the main output inductor LI for a single-ended forward converter and filter, as shown in Fig. 1.20Ja. The specification for the converter is as follows The design approach will assume that the output ripple current must not exceed 30 of I, (6 A p-p in this example). Also, to allow for a range of control, the pulse width at nominal input will be 30 of the total period (that is 10 jxs). To provide an output of 5 Vat a pulse width of 30 , the transformer...

2012 Output Capacitor Value

It is normally assumed that the output capacitor size will be determined by the ripple current and ripple voltage specifications only. However, if a second-stage output filter L2, C2 is used, a much higher ripple voltage could be tolerated at the terminals of Cl without compromising the output ripple specification. Hence if ripple voltage were the only criterion , a much smaller capacitor could be used. For example, assume that the ripple voltage at the terminals of Clean be 500 mV. The current...

202 Basic Requirements

The following section on output-filter design assumes that normal good design practice has already been applied to minimize conducted-mode noise and that RFI filters have been fitted to the input supply lines, as specified in Sec. 3.1. To provide a steady DC output, and reduce ripple and noise, LC low-pass filters (as shown in Fig. 1.20. la) will normally be provided on switching supply outputs. In forward converters, these filters carry out two main functions. The prime requirement is one of...

2022 Type 2 Boost Regulators

Figure 2,20.2a shows the general arrangement of the power sections of a boost regulator. The operation is as follows. When Ql turns on, the supply voltage will be impressed across the series inductor LI. Under steady-state conditions, the current in LI will increase linearly in the forward direction. Rectifier D1 will be reverse-biased and not conducting. At the same time (under steady-state conditions), current will be flowing from the output capacitor CI into the load. Hence, CI will be...

203 Control And Drive Circuits

There are many suitable control and drive circuits in both discrete and integrated circuit form. Many of the single-ended control circuits used for the forward and flyback converters can be used for the Cuk regulator. Although many switching regulator control circuits use duty ratio control quite successfully, the more recent current-mode control techniques can be applied. These will yield advantages similar to those found in conventional transformer converters. Research and development work is...

203 Parasitic Effects In Switchmode Output Filters

Figure 1.20.1a shows a single-stage LC output filter (such as might be found in a typical forward converter. It includes the parasitic elements Cc, R ESL, and ESR. The series inductor arm LI shows an ideal inductor L in series with the inevitable winding resistance R. The parasitic distributed interwinding capacitance is included as lumped equivalent capacitor Cc, The shunt capacitor CI includes the effective series inductance ESL and the effective series resistance ESR. The equivalent circuit...

204 Inductor Design For Switching Regulators

It will be clear from the preceding that the inductors (or chokes) play a critical part in the performance of the regulators. These inductors carry a large component of DC current, as well as sustaining a large high-frequency ac stress. The inductor must not saturate for any normal condition, and for good efficiency the winding and core losses must be small. The choice of inductance is usually a compromise. Theoretically, the inductance can have any value. Large values are expensive and lossy,...

204 Twostage Filters

As shown above, attempts to satisfy all the voltage averaging and noise rejection requirements in a single LC filter would require the selection of expensive components, particularly in flyback converters. Even then, only mediocre high-frequency performance would be obtained. Figure 1.20.2 shows how a far more cost-effective wideband filter can be produced, using a second-stage, much smaller, LC filter to reject the high-frequency noise. The second stage L2, C2, may be quite small and...

205 Highfrequency Choke Example

To get the best performance from the high-frequency choke L2, the interwinding capacitance should be minimized. Figure 1.20.3a shows a 1-in-long ferrite rod choke with a yi6-in diameter, wound with 15 turns of closely packed 17 AWG wire. Figure 1.20,36 shows a plot of phase shift and impedance as a function of frequency for this choke. The phase shift is zero at the self-resonant frequency, which in this case is 4.5 MHz. The impedance plot in Fig. 1.20.3c shows the improvement obtained by...

205 Inductor Design Example

Calculate the inductance required for a 10-A, 5-V type 1 buck regulator operating at 40 kHz with an input voltage from 10 to 30 V, when the ripple current is not to exceed 20 of IDC (2 A). Procedure Maximum ripple current will occur when the input voltage is maximum that is, when the voltage applied across the inductor is maximum. 1. Calculate the on time when the input is 30 V. where tp total period (ton + foff) 2. Select the peak-to-peak ripple current. This is by choice 20 of DC, or 2 A in...

206 Resonant Filters

By selecting capacitors such that their self-resonantfrequency is near the switching frequency, the best performance will be obtained. Many of the small, low-ESR electrolytic capacitors have a series self-resonant frequency near the typical operating frequencies of switchmode converters. At the self-resonantfrequency, the parasitic internal inductance of the capacitor resonates with the effective capacitance to form a series resonant circuit. At this frequency, the capacitor impedance tends to...

207 Resonant Filter Example

Figure 1.20.5 shows a typical output stage of a small 30-kHz, 5-V, 10-A flyback converter with a two-stage output filter. (In flyback converters, the transformer inductance and CI form the first stage of the J.C power filter.) A second stage high-frequency filter L2, C2 has been added. For this example, the same 1 in, 5 i6-in-diameter ferrite rod inductor used to obtain plot c in Fig. 1.20.3 is used for L2. The 15 spaced turns on this rod give an inductance of 10 fi.H and a low interwinding...

207 The Ripple Regula

A control technique which tends to be reserved for the buck-type switching regulator is the so-called ripple regulator.17 This is worthy of consideration here, as it provides excellent performance at very low cost. The ripple regulator is best understood by considering the circuit of the buck regulator shown in Fig. 2.20,5a. A high-gain comparator amplifier A1 compares a fraction of the output voltage Vou, with the reference VR when the output fraction is higher than the reference, the series...

208 Commonmode Noise Filters

The discussion so far has been confined to series-mode conducted noise. The f j. ter described will not be effective for common-mode noise, that is, noise voltages appearing between the output lines and the ground plane. The common-mode noise component is caused by capacitive or inductive coupling between the power circuits and the ground plane within the p & supply. Initially this must be reduced to a minimum by correct screening and layout at the design stage. Further reduction of the...

2110 Some Advantages Of The Saturable Reactor Regulator

For low-voltage, high-current secondary outputs, the saturable reactgr control is particularly valuable. The onv-state impedance may be very close to the resistance of the copper winding (afew milliohms in high-current applications). Consequently, the voltage drop across the reactor element will be very low in the on state. In the off state, with the right core material, the inductance and hence the reactance can be very high, and leakage current (magnetization current) is low. Consequently,...

2111 Some Limiting Factors In Saturable Reactor Regulators

The saturable reactor is not a perfect switch. Several obvious limitations, such as maximum and minimum off' and on reactance, have already been mentioned. Some of the less obvious but important limitations will now be considered. I. Parasitic Reset. When the voltage applied to the reactor reverses during the reset period, the main rectifier diode D1 must block (turn off). During this blocking period, there will be a reverse recovery current flowing in the diode. This reverse current flows in...

2112 The Case For Constantvoltage Or Constantcurrent Reset Highfrequency Instability Considerations

At high frequencies the area of the BIH loop increases, giving an increased core loss and a general degradation of the desirable magnetic properties. In particular, some materials show a modification of the BIH loop to a pronounced S-shaped characteristic. This S shape can lead to instability if constant-current resetting is used in the control circuit. This effect is best understood by considering Fig. 2.21.10. If constant-current reset is used, then the magnetizing force H is the controlled...

21131 Core Material

The choice of core material is normally a compromise between cost and performance. At low frequencies, there are many suitable materials, and the controlling factors will be squareness ratio, saturation flux density, cost, and core losses. Also, at low frequencies, core losses are less important, giving a wider selection. At medium frequencies, up to, say, 35 kHz, the core loss starts to be the predominant factor, and Permalloys, square ferrites, or amorphous materials will be chosen. At high...

212 Operating Principles

In simple terms, the saturable reactor is. used in high-frequency switchmode supplies as a flux-saturation-controlled power switch, providing regulation by secondary pulse-width control techniques. The method of operation is best explained by considering the conventional buck regul'*''- circuit shown in Fig. 2.21.1. This figure shows the output LC filter and rectif such as would be found on the secondary of a typical single-ended FIG. 2.21.1 . Typical secondary output rectifier and filter cir-...

213 Simple Power Failure Warning Circuits

Figure 1.21.1 shows a simple optically coupled circuit typical of those often used for power failure warning. However, it will be shown that this type of circuit is suitableonly for type 1 failures, that is, totallinefailureconditions.lt operates as follows. The ac line input is applied to the network R1 and bridge rectifier D1 such that unidirectional current pulses flow in the optical coupler diode. This maintains a pulsating conduction of the optical coupler transistor Ql. While this...

214 Dynamic Power Failure Warning Circuits

The more complex dynamic power failure warning circuits are able to respond to brownout conditions. Many types of circuit are in use, and it may be useful to examine some of the advantages and disadvantages of some of the more common techniques. Figures 1.21.2 and 1.21.3 show two circuits that will ensure that sufficient warning of failure is given for all conditions. In the first example, a fraction of the DC voltage on the power converter reservoir capacitors CI and C2 is compared with a...

215 Independent Power Failure Warning Module

The previous two power failure circuits must be part of the power supply, as they depend on the internal DC header voltage for their operation. Figure 1.21.4 shows a circuit that will operate directly from the line input and is independent of Ihis circuit has its own bridge rectifier D1-D4, which again provides a unidirectional half-sine-wave input to the feed resistor Rl, ZD1, and the optical cou- EE3. 1.21.4 Independent power failure module for direct operation from ac line inputs. EE3....

215 Saturable Reactor Quality Factors

The effectiveness of the saturable reactor as a power switch will be determined by several factors as follows The magnetization current can be considered a leakage current in the off' state of the switch. The reactor's quality as an off' switch that is, its maximum impedance_ will be defined by its maximum inductance. This in turn depends on the permeability of the core in the unsaturated state and the number of turns. Increasing the number of turns will, of course, increase the inductance and...

216 Power Failure Warning In Flyback Converters

Very simple power failure warning circuits can be fitted to flyback converters, because in the forward direction the flyback transformer is a true transformer, providing an isolated and transformed output voltage which is proportional to the applied DC. Figure 1.21.5 shows the power section of a simple single-output flyback supply providing a 5-V output. Diode D1 conducts in the flyback mode of T1 to charge C2 and deliver the required 5-V output. The control circuit adjusts the duty cycle in...

217 Controlling The Saturable Reactor

As explained in Sec. 21.2, to control the saturable reactor in switching regulator applications, it is necessary to reset the core during the off' period to a defined position on the BIH characteristic prior to the next forward power pulse. The reset (volt-seconds), may be applied by a separate control winding (transductor or magnetic amplifier operation see Fig. 2.21.7) or by using the same primary power winding and applying a reset voltage in the opposite direction to the previous power pulse...

217 Fast Power Failure Warning Circuits

The previous systems shown in this section respond quite slowly to brownout conditions, because they are sensing peak or mean voltages. The filter capacitor in the warning circuit introduces a delay. Its value is a compromise, being low enough to prevent a race between the holdup time of the power supply and the time constant of the filter capacitor, but large enough to give acceptable ripple voltage reduction. It is possible to detect the imminent failure of the line before this has fully...

218 Current Limiting The Saturable Reactor Regulator

Figure 2.21.8 shows a simple current-limiting circuit which operates as follows. When the output current is such that the voltage across RI exceeds 0.6 V -< n- FIG. 2.21.8 Saturable reactor buck regulator with current-limiting circuit R1 and Q2. FIG. 2.21.8 Saturable reactor buck regulator with current-limiting circuit R1 and Q2. sistor Q2 will conduct and provide a reset current via D2 when the secondary voltage goes negative, thus limiting the maximum current. When this type of current...

219 Pushpull Saturable Reactor Secondary Power Control Circuit

The discussion so far has been limited to single-ended systems. In such systems, the same time (volt-seconds) is required to reset the core during the off period as was applied to the core to set it during the on period. Therefore, if control is to be maintained under short-circuit conditions, the duty ratio cannot exceed 50 unless a high-voltage reset circuit is provided or a reset tapping point is provided on the SR winding. In the push-pull system shown in Fig. 2.21.9, two saturable reactors...

221 Introduction

Most engineers will be very familiar with the general performance parameters of constant-voltage power supplies. They will recognize that these power supplies have a limited power capability, normally with fixed output voltages and some form of current- or power-limited protection. For example, a 10-V 10-A power supply would be expected to deliver from zero to 10 A at a constant output voltage of 10 V. Should the load current try to exceed 10 A, the supply would be expected to limit the...

222 Constantvoltage Suppues

From Fig. 2.22.1, the output characteristics of the constant-voltage supply will be recognized. The normal working range for the constant-voltage supply will be for load resistances from infinity (open circuit) to 1 R. In this range, the load current is 10 A or less. The voltage is maintained constant at 10 V in this working range. At load resistances of less than 1 R, the current-limited area of operation will be entered. In a constant-voltage supply, this is recognized as an overload...

222 The Effect of an Air Gap on the AC Conditions

It is clear from Fig. 2.2.1 b that increasing the core gap results in a decrease in the slope of thpfBi characteristic but does not change the required AEi Hence there is an increase in the magnetizing current iHt This corresponds to an effective reduction in the permeability of the core and a reduced primary inductance. Hence, a core gap does not change the ac flux density requirements or otherwise improve the ac performance of the core. A common misconception is to assume that a core which is...

223 Saturable Reactor Voltage Adjustment

Consider the effect of placing a saturable reactor toroid (as described in Part 2, Chap. 21) on the output lines from the transformer to the 12-V rectifiers D1 and D2. These reactors LI and L2 are selected and designed so that they take a time-delay period td to saturate, specified by required VolU X ion U 'on actual Vout 1X5 The extra time delay td is introduced on the leading edge of the output p0wer pulse by the saturable reactor, and the 12.7-V output would be adjusted back to 12 V. This...

223 The Effect of an Air Gap on the DC Conditions

A DC current component in the windings gives rise to a DC magnetizing force Hjx on the horizontal H axis of the BIHloop, ( dc is proportional to the mean DC ampere-turns.) For a defined secondary current loading, the value of DC is defined. Hence, for the DC conditions, B may be considered the dependent variable. It should be noted that the gapped core can support a much larger value of H (DC current) without saturation. Clearly, the higher value of H, HDC2, would be sufficient to saturate the...

225 Problems

What is meant by the term centering as applied to multiple-output converters 2. Why is centering sometimes required in multiple-output applications 3. Describe a method of nondissipative voltage centering commonly used in ratio-controlled converters. 4. Explain how saturable reactors LI and L2 in Fig. 1.22.1 reduce the output voltages of the 12-V outputs. 5. Assume that the single-ended forward converter shown in Fig. 1.22.1 gives the required 5-V output when the duty ratio is 40 at a frequency...

23 General Design Considerations

In the following design, the ac and DC conditions applied to the primary are dealt with separately. Using this approach, it will be clear that the applied ac voltage, frequency, area of core, and maximum flux density of the core material control the minimum primary turns, irrespective of core permeability, gap size, DC current, or required inductance. It should be noted that the primary inductance will not be considered as a transformer design parameter in the initial stages. The reason for...

231 Introduction

Very often, an auxiliary power supply will be required, to provide power for control and drive circuits within the main switchmode unit. Depending on the chosen design approach, the auxiliary supply will be common to either input or output lines, or in some cases will be completely isolated. A number of ways of meeting these auxiliary requirements are outlined in the following sections. The method chosen to provide the auxiliary needs should be considered very carefully, as this choice will...

232 60hz Line Transformers

Very often small 60-Hz transformers will be used to develop the required auxiliary power. Although this may be convenient, as it allows the auxiliary circuits to be energized before the main converter, the 60-Hz transformer tends to be rather large, as it must be designed to meet the insulation and creepage requirements of the various safety specifications. Hence, the size, cost, and weight of a 60-Hz auxiliary supply transformer tends to make it less attractive for the smaller switchmode...

233 Auxiliary Converters

Very small, lightweight auxiliary power supplies can be made using self-oscillating high-frequency flyback converters. The output windings on the converter can be completely isolated and provide both input and output auxiliary needs, in the same way as the previous 60-Hz transformers. Because auxiliary power requirements are usually very small (5 W or less), extremely small and simple converters can be used. A typical example of a nonregulated auxiliary converter is shown in Fig. 1.23.1. In...

233 Drive Circuit

Figure 2.23.2 shows the basic elements of the drive circuit for the cascaded power sections. VH*+70Y Rl 'C Ql +Y OUT 0-60V Figure 2.23.2 shows the basic elements of the drive circuit for the cascaded power sections. VH*+70Y Rl 'C Ql +Y OUT 0-60V The operation is best understood by considering four extreme operating conditions. These will be 1. Low-output-voltage, high-current conditions 2. High-output-voltage, high-current conditions 3. Intermediate-voltage, high-current conditions (say, 30 V,...

234 Operating Principles

Initially, Ql starts to turn on as a result of the base drive current in resistors Rl and R2. As soon as Ql starts to turn on, regenerative feedback via winding P2 will assist the turn-on action bf the transistor, which will now latch to an on state. With Ql on, current will build up linearly in the primary winding at a rate defined by the primary inductance and applied voltage dI dt< xVcc LP). As the current builds up in the collector and emitter of Ql, the voltage across R3 will crease. The...