102 Operating Principles

FET 1 and FET 2 (the power switches) turn on, or off, simultaneously. When the devices are switched on, the primary supply voltage Vci will be applied across the transformer primary, and the starts of all windings will go positive. Under steady-state conditions, a current will have been established in the output choke LI by previous cycles, and this current will be circulating by flywheel action in the choke LI, capacitor CI, and load, returning via the flywheel diode When the secondary emf is...

1101

EES. 1.12.1 Characteristics of a (ypical undervoltage transient protection circuit. (a) Load current transient. (b) Typical undervoltage transient excursion without protection circuit fitted, (c) Undervoltage transient excursion with protection circuit fitted. EES. 1.12.1 Characteristics of a (ypical undervoltage transient protection circuit. (a) Load current transient. (b) Typical undervoltage transient excursion without protection circuit fitted, (c) Undervoltage transient excursion with...

111 Specification Example For A 110w Directoffline Flyback Power Supply

For the following example, a fixed-frequency single-ended bipolar flyback unit with three outputs and a power of 110 W is to be considered. It will be shown later that the same design approach is applicable to variable-frequency self-oscillating units. Although most classical design approaches assume that the mode of operation will be either entirely complete energy transfer (discontinuous mode) or entirely incomplete energy transfer (continuous mode), in practice a system is unlikely to remain...

1113 Transformer Design

The design of the transformer for this power supply is shown in Part 2, Chap. 2. 1.72 PROBLEMS 1. From what family of converters is the flyback converter derived 2. During which phase of operation is the energy transferred to the secondary in a flyback converter . 3. Describe the major advantages of the flyback technique. 4. Describe the major disadvantages of the flyback technique. 5. Why is the transformer utilization factor of a flyback converter often much lower than that of a push-pull...

114 Crowbar Performance

More precise crowbar protection circuits are shown in Fig. 1.11.1b and c. The type of circuit selected depends on the performance required. In the simple crowbar, there is always a compromise choice to be made between ideal fast protection (with its tendency toward nuisance operation) and delayed operation (with its potential for voltage overshoot during the delay period). For optimum protection, a fast-acting, nondelayed overvoltage crowbar is required. This should have an actuation voltage...

115 Limitations Of Simple Crowbar Circuits

The well-known simple crowbar circuit shown in Fig. 1.11. la is popular for many noncritical applications. Although this circuit has the advantages of low cost and circuit simplicity, it has an illdefined operating voltage, which can cause large operating spreads. It is sensitive to component parameters, such as temperature coefficient and tolerance spreads in the zener diode, and variations in the gate-cathode operating voltage of the SCR. Furthermore, the delay time provided by Cl is also...

116 Type 2 Overvoltage Clamping Techniques

FIG. 1.11.2 Shunt regulator-type voltage damp circuits. is highly dissipative, and the source resistance must limit the current to acceptable levels. Hence, shunt clamping action can be used only where the source resistance (under failure conditions) is well specified and large. In many cases shunt protection of this type relies on the action of a separate current or power limiting circuit for its protective performance. An advantage of the clamp technique is that there is no delay in the...

12 Expected Performance

In the example shown in Fig. 2.1.1, the main output is closed-loop-controlled and is thus fully regulated. The auxiliary outputs are only semiregulated and may be expected to provide line and load regulation of the order of 6 . Where better regulation is required, additional secondary regulators will be needed. In flyback supplies, secondary regulators are often linear dissipative types, although switching regulators may be used for higher efficiency. For low-current outputs, the standard...

121 Output Ripple and Noise

Where very low levels of output ripple are required, the addition of a small LC noise filter near the output terminals will often eliminate the need for expensive low-ESR capacitors in the main secondary reservoir positions. For example, a typical 5-V 10-A supply may use the highest-quality low-ESR capacitors in positions Ci, C2, and C3 of the single-stage filter shown in Fig. 2.1.1, but this will rarely give a ripple figure of less than 100 mV. However, it is relatively easy to keep ripple...

1210 Optimum Flux Density

The choice of optimum flux density 5opt will be a matter for careful consideration. Unlike with the flyback converter, both quadrants of the BIH loop will be used, and the available induction excursion is more than double that of the flyback case. Consequently, core losses are to be considered more carefully as these may exceed the copper losses if the full induction excursion is used. For the most efficient design, the copper and core losses should be approximately equal. Figure 2.12.2 shows...

1212 Calcula Ting Primary Turns

Once optimum core size and peak flux density have been selected, the primary turns may be calculated. The transformer must provide full output voltage at minimumline input. Under these conditions, the power pulse will have its maximum width of 16.5 (jls. Hence the minimum primary turns are calculated for this condition. With 90 V rms input to the voltage doubling network, the DC voltage will be approximately 222 V. (See Part 1, Chap. 6.) Consider one half cycle of operation. The capacitors CI...

1215 Control And Drive Circuits

The control and drive circuits used for this type of converter are legion. They range from fully integrated control circuits, available from a number or manufacturers, to the fully discrete designs favored by many power supply engineers. A discussion of suitable drie circuits will be found in Part 1, Chaps. 15 and 16. For reliable operation, the drive and control circuits must provide the following basic functions 1. Soft start. This reduces inrush current and turn-on stress, and helps to...

1216 Flux Doubling Effect

The difference in the operating mode for'the single-ended transformer and the push-pull balanced transformer is not always fully appreciated. For the single-ended forward or flyback converter, only one quadrant of the BIH loop is used, there is a remnant flux B and the remaining range of induction is often quite small. (Figure 2.9,2a and b shows the effect well.) in the push-pull transformer, it is normally assumed that the full BIH loop may be used and that B will be incremented from Bmto each...

122 Operating Principles

Figure 2.12,1 a shows the general arrangement of power sections for the half-bridge push-pull converter. The switching transistors Ql and Q2 form only one side of the bridge-connected circuit, the remaining half being formed by the two capacitors Cl and C2. The major difference between this and the full bridge is that the primary of the transformer will see only half the supply voltage, and hence the current in the winding and switching transistors will be twice that in the full-bridge case....

122 Undervoltage Suppressor Performance Parameters

Figure 1.12. la, b, and c shows the typical current and voltage waveforms-that may be expected at the load end of the DC output lines from a power supply when a large transient load is applied by the load. Figure 1.12.1a shows a large transient load current demand during the period from t, to t,. Figure 1.12.1& shows the undervoltage transient that might typically be expected at the load during this transient. (Assume that the voltage dip is caused by the resistance and inductance of the...

124 Practical Circuit Description

Figure 1.12.4 shows a practical implementation of this technique. In this circuit, switch SW1 or Q1 is replaced by Darlington-connected transistors Q3 and Q4. These transistors operate as a switch and linear regulator. FIG. 1.12.4 Example of an undervoltageprotection circuit. FIG. 1.12.4 Example of an undervoltageprotection circuit. Although Q3 and Q4 are now shown positioned between the two capacitors CI and C2, it has been demonstrated above that since they still form a series network, their...

124 Problem Areas

The designer must guard against a number of possible problems with this type of converter. A major difficulty is staircase saturation of the transformer core. If the average volt-seconds applied to the primary winding for all positive-going pulses is not exactly equal to that for all negative-going pulses, the transformer flux density will increase with each cycle (staircase) into saturation. The same effect will occur if the secondary diode voltages are unbalanced. As storage times and...

13 Operating Modes

Two modes of operation are clearly identifiable in the flyback converter 1. Complete energy transfer (discontinuous mode), in which all the energy that was stored in the transformer during an energy storage period (on period) is transferred to the output during the flyback period (off period). 2. Incomplete energy transfer (continuous mode), in which a part of the en-ergy scored in the transformer at the end of an on period remains in the transfohner at the beginning of the next on period.

131 Transfer Function

The small-signal transfer functions for these two operating modes are quite different, and they are dealt with separately in this section. In practice, when a wide range of input voltages, output voltages, and load currents is required, the flyback converter will be required to operate (and be stable) in both complete and incomplete energy transfer modes, since both modes will be encountered at some point in the operating range. As a result of the change in transfer function at the point where...

1321 General Conditions

Figure 2.13.1 shows the power section of a typical off-line bridge converter. Diagonal pairs of switching devices are operated simultaneously and in alternate sequence. For example, Ql and Q3 would both be on at the same time, followed by Q2 and Q4. In a pulse-width-controlledsystem, there will be a period when all four devices will be off. It should be noted that when Q2 and Q4 are on, the voltage across the primary winding has been reversed from that when Ql and Q3 were on. In this example, a...

133 Type 1 Overpower Limiting

The first type is a power-limiting protection method, often used in flyback units or supplies with a single output. It is primarily a power supply short-circuit protection technique. This and the methods used in types 2 and 4 are electronic, and depend on the power supply remaining in a serviceable condition. The supply may be designed to shut down or self-reset if the overload is removed. In this type of protection, the power (usually in the primary side of the con- verter transformer) is...

134 Type 1 Forma Primary Overpower Limiting

In this form of power limiting, the primary power is constantly monitored. If the load tries to exceed a defined maximum, the input power is limited to prevent any further increase. Usually, the shape of the output current shutdown characteristic is poorly defined when primary power limiting is used on its own. However, because of its low cost, primary power limiting has become generally accepted in lower-power, low-cost units (particularly in multi-output flyback power supplies). It should be...

135 Type 1 Form B Delayed Overpower Shutdown Protection

One of the most effective overload protection methods for low-power, low-cost supplies is the delayed overpower shutdown technique. This operates in such a way that if the load power exceeds a predetermined maximum for a duration beyond a short defined safe period, the power supply will turn off, and an input power off-on cycle will be required to reset it to normal operation. Not only does this technique give the maximum protection to both power supply and load, but it is also the most...

136 Type 1 Form C Pulsebypulse Overpowercurrent Limiting

This is a particularly useful protection technique that will often be used in addition to any secondary current limit protection. * The input current in the primary switching devices is monitored on a real-time basis. If the current exceeds a defined limit, the on pulse is terminated. With discontinuous flyback units, the peak primary current defines the power, and hence this type of protection becomes a true power limit for such units. With the forward converter, the input power is a function...

142 Foldback Principle

Figure 1.14.1 shows a typical reentrant characteristic, as would be developed measured at the output terminals of a foldback-limited power supply. A purely resistive load will develop a straight load line (for example, the 5-fl load line shown in Fig. 1.14.1). A resistive load line has its point of origin at zero, and the current is proportional to voltage. As a resistive load changes, the straight line (which vill start vertically at zero load i.e., infinite resistance) will swing clockwise...

143 Foldback Circuit Principles As Applied To A Linear Supply

FOLDBACK output CURRENT LIMITING 1.115 14. FOLDBACK output CURRENT LIMITING 1.115 REGULATOR TRANSISTOR AND CURRENT LIMIT CIRCUIT REGULATOR TRANSISTOR AND CURRENT LIMIT CIRCUIT ED3. 1.14.2 (a) Foldback current limit circuit, (b) Regulator dissipation with reentrant protection. ED3. 1.14.2 (a) Foldback current limit circuit, (b) Regulator dissipation with reentrant protection. put voltage is zero (output short circuit). At short circuit, the current in R1 is very small, and the voltage across...

144 Lockout In Foldback Currentlimited Suppues

With the resistive load (the straight-line loads depicted in Figs. 1.14.1 and 1.14.3), there can only be one stable point of operation, defined by the intersection of the NON LINEAR LOAD LINE (LOCK OUT AT 'P2') EK3. 1.14.3 Overload and start-up characteristics of a foldback, current-limited supply, showing performance for linear and nonlinear load lines. NON LINEAR LOAD LINE (LOCK OUT AT 'P2') EK3. 1.14.3 Overload and start-up characteristics of a foldback, current-limited supply, showing...

145 Reentrant Lockout With Crossconnected Loads

Lockout problems can occur even with linear resistive loads when two or more foldback-limited power supplies are connected in series. (This series connection is often used to provide a positive and negative output voltage with respect to a common line.) In some cases series power supplies are used to provide higher output voltages. Figure 1.14,5a shows a series arrangement of foldback-limited supplies. Here, positive and negative 12-V outputs are provided. The normal resistive loads R1 and R2...

15 Energy Storage Phase

The energy storage phase is best understood by considering the action of the basic single-output flyback converter shown in Fig. 2.L2. When transistor Ql is turned on, the start of all windings on the transformer FIG. 2.1.2 Simplified power section of a flyback (buck-boost) converter. will go positive. The output rectifier diode D1 will be reverse-biased and will not conduct therefore current will not flow in the secondary while Q1 is conducting. During this energy storage phase only the...

16 Energy Transfer Modes Flyback Phase

When Q1 turns off, the primary current must drop to zero. The transformer ampere-turns cannot change without a corresponding change in the flux density LB. As the change in the flux density is now negative-going, the voltages will reverse on all windings (flyback action). The secondary rectifier diode Dl will conduct, and the magnetizing current will now transfer to the secondary. It will continue to flow from start to finish in the secondary winding. Hence, the set-ondary (flyback) current...

166 Widerange Proportional Drive Circuits

Where the range of input voltage and load are very wide, the circuit shown in Fig. 1.16.1 will have some limitations, as follows. When the input voltage is low, the duty-cycle will be large, and Ql may be on for periods considerably exceeding 50 of the total period. Further, if the minimum load is small, LI will be large to maintain continuous conduction in the output filter. Under these conditions, the collector current is small, but the on period is long. During the long on period, a...

1810 The Wea Ving Lowloss Snubber Diode

As shown above, to reduce secondary breakdown stress during the turn-off of high-voltage bipolar transistors, it is normal practice to use a snubber network. Unfortunately, in normal snubber circuits, a compromise choice must be made between a high-resistance snubber (to ensure a low turn-on current) and a low-resistance snubber (to prevent a race condition at light loads where narrow pulse widths require a low CR time constant). This paradox often results in a barely satisfactory compromise....

182 Snubber Circuit With Load Line Shaping

Figure 1.18. la shows the primary of a conventional single-ended flyback converter circuit PI, Q1 with a leakage inductance energy recovery winding and diode P2, D3. Snubber components Dl, Cl, and R1 are fitted from the collector to the emitter of Ql. Figure 1.18,16 shows the voltage and current waveforms to be expected in this circuit. If load line shaping is required, then the main function of the snubber components is to provide an alternative path for the inductively maintained primary...

188 Dissipation In Snubber Resistor

The energy dissipated in the snubber resistor during each cycle is the same as the energy stored in Cl at the end of the off' period. However, the voltage across Cl depends on the type of converter circuit. With complete energy transfer, the voltage on Cl will be the supply voltage Vcc, as all flyback voltages will have fallen to zero before the next on period. With continuous-mode operation, the voltage will be the supply voltage plus the reflected secondary voltage. Having established the...

194 Circuit Operation

Consider Figs. 1.19.2 and 1.19.3 for the initial condition when Q2 is just about to turn on (point tl in the drive waveform). At this instant, input 1 of gate U3 is enabled for an on state of Q2. However, as a result of its storage time, Ql will still be conducting and its collector voltage will be low. Hence, input 2 of U3 will be low. As a result of the gating action of U3, the turn-on of Q2 is delayed until the voltage on the collector of Ql goes high. This does not occur until the end of...

2012 Output Capacitor Value

It is normally assumed that the output capacitor size will be determined by the ripple current and ripple voltage specifications only. However, if a second-stage output filter L2, C2 is used, a much higher ripple voltage could be tolerated at the terminals of Cl without compromising the output ripple specification. Hence if ripple voltage were the only criterion , a much smaller capacitor could be used. For example, assume that the ripple voltage at the terminals of Clean be 500 mV. The current...

2022 Type 2 Boost Regulators

Figure 2,20.2a shows the general arrangement of the power sections of a boost regulator. The operation is as follows. When Ql turns on, the supply voltage will be impressed across the series inductor LI. Under steady-state conditions, the current in LI will increase linearly in the forward direction. Rectifier D1 will be reverse-biased and not conducting. At the same time (under steady-state conditions), current will be flowing from the output capacitor CI into the load. Hence, CI will be...

205 Inductor Design Example

Calculate the inductance required for a 10-A, 5-V type 1 buck regulator operating at 40 kHz with an input voltage from 10 to 30 V, when the ripple current is not to exceed 20 of IDC (2 A). Procedure Maximum ripple current will occur when the input voltage is maximum that is, when the voltage applied across the inductor is maximum. 1. Calculate the on time when the input is 30 V. where tp total period (ton + foff) 2. Select the peak-to-peak ripple current. This is by choice 20 of DC, or 2 A in...

207 Resonant Filter Example

Figure 1.20.5 shows a typical output stage of a small 30-kHz, 5-V, 10-A flyback converter with a two-stage output filter. (In flyback converters, the transformer inductance and CI form the first stage of the J.C power filter.) A second stage high-frequency filter L2, C2 has been added. For this example, the same 1 in, 5 i6-in-diameter ferrite rod inductor used to obtain plot c in Fig. 1.20.3 is used for L2. The 15 spaced turns on this rod give an inductance of 10 fi.H and a low interwinding...

207 The Ripple Regula

A control technique which tends to be reserved for the buck-type switching regulator is the so-called ripple regulator.17 This is worthy of consideration here, as it provides excellent performance at very low cost. The ripple regulator is best understood by considering the circuit of the buck regulator shown in Fig. 2.20,5a. A high-gain comparator amplifier A1 compares a fraction of the output voltage Vou, with the reference VR when the output fraction is higher than the reference, the series...

208 Commonmode Noise Filters

The discussion so far has been confined to series-mode conducted noise. The f j. ter described will not be effective for common-mode noise, that is, noise voltages appearing between the output lines and the ground plane. The common-mode noise component is caused by capacitive or inductive coupling between the power circuits and the ground plane within the p & supply. Initially this must be reduced to a minimum by correct screening and layout at the design stage. Further reduction of the...

212 Operating Principles

In simple terms, the saturable reactor is. used in high-frequency switchmode supplies as a flux-saturation-controlled power switch, providing regulation by secondary pulse-width control techniques. The method of operation is best explained by considering the conventional buck regul'*''- circuit shown in Fig. 2.21.1. This figure shows the output LC filter and rectif such as would be found on the secondary of a typical single-ended FIG. 2.21.1 . Typical secondary output rectifier and filter cir-...

213 Simple Power Failure Warning Circuits

Figure 1.21.1 shows a simple optically coupled circuit typical of those often used for power failure warning. However, it will be shown that this type of circuit is suitableonly for type 1 failures, that is, totallinefailureconditions.lt operates as follows. The ac line input is applied to the network R1 and bridge rectifier D1 such that unidirectional current pulses flow in the optical coupler diode. This maintains a pulsating conduction of the optical coupler transistor Ql. While this...

214 Dynamic Power Failure Warning Circuits

The more complex dynamic power failure warning circuits are able to respond to brownout conditions. Many types of circuit are in use, and it may be useful to examine some of the advantages and disadvantages of some of the more common techniques. Figures 1.21.2 and 1.21.3 show two circuits that will ensure that sufficient warning of failure is given for all conditions. In the first example, a fraction of the DC voltage on the power converter reservoir capacitors CI and C2 is compared with a...

216 Power Failure Warning In Flyback Converters

Very simple power failure warning circuits can be fitted to flyback converters, because in the forward direction the flyback transformer is a true transformer, providing an isolated and transformed output voltage which is proportional to the applied DC. Figure 1.21.5 shows the power section of a simple single-output flyback supply providing a 5-V output. Diode D1 conducts in the flyback mode of T1 to charge C2 and deliver the required 5-V output. The control circuit adjusts the duty cycle in...

217 Controlling The Saturable Reactor

As explained in Sec. 21.2, to control the saturable reactor in switching regulator applications, it is necessary to reset the core during the off' period to a defined position on the BIH characteristic prior to the next forward power pulse. The reset (volt-seconds), may be applied by a separate control winding (transductor or magnetic amplifier operation see Fig. 2.21.7) or by using the same primary power winding and applying a reset voltage in the opposite direction to the previous power pulse...

219 Pushpull Saturable Reactor Secondary Power Control Circuit

The discussion so far has been limited to single-ended systems. In such systems, the same time (volt-seconds) is required to reset the core during the off period as was applied to the core to set it during the on period. Therefore, if control is to be maintained under short-circuit conditions, the duty ratio cannot exceed 50 unless a high-voltage reset circuit is provided or a reset tapping point is provided on the SR winding. In the push-pull system shown in Fig. 2.21.9, two saturable reactors...

222 The Effect of an Air Gap on the AC Conditions

It is clear from Fig. 2.2.1 b that increasing the core gap results in a decrease in the slope of thpfBi characteristic but does not change the required AEi Hence there is an increase in the magnetizing current iHt This corresponds to an effective reduction in the permeability of the core and a reduced primary inductance. Hence, a core gap does not change the ac flux density requirements or otherwise improve the ac performance of the core. A common misconception is to assume that a core which is...

23 General Design Considerations

In the following design, the ac and DC conditions applied to the primary are dealt with separately. Using this approach, it will be clear that the applied ac voltage, frequency, area of core, and maximum flux density of the core material control the minimum primary turns, irrespective of core permeability, gap size, DC current, or required inductance. It should be noted that the primary inductance will not be considered as a transformer design parameter in the initial stages. The reason for...

232 60hz Line Transformers

Very often small 60-Hz transformers will be used to develop the required auxiliary power. Although this may be convenient, as it allows the auxiliary circuits to be energized before the main converter, the 60-Hz transformer tends to be rather large, as it must be designed to meet the insulation and creepage requirements of the various safety specifications. Hence, the size, cost, and weight of a 60-Hz auxiliary supply transformer tends to make it less attractive for the smaller switchmode...

233 Auxiliary Converters

Very small, lightweight auxiliary power supplies can be made using self-oscillating high-frequency flyback converters. The output windings on the converter can be completely isolated and provide both input and output auxiliary needs, in the same way as the previous 60-Hz transformers. Because auxiliary power requirements are usually very small (5 W or less), extremely small and simple converters can be used. A typical example of a nonregulated auxiliary converter is shown in Fig. 1.23.1. In...

233 Drive Circuit

Figure 2.23.2 shows the basic elements of the drive circuit for the cascaded power sections. VH*+70Y Rl 'C Ql +Y OUT 0-60V Figure 2.23.2 shows the basic elements of the drive circuit for the cascaded power sections. VH*+70Y Rl 'C Ql +Y OUT 0-60V The operation is best understood by considering four extreme operating conditions. These will be 1. Low-output-voltage, high-current conditions 2. High-output-voltage, high-current conditions 3. Intermediate-voltage, high-current conditions (say, 30 V,...

234 Operating Principles

Initially, Ql starts to turn on as a result of the base drive current in resistors Rl and R2. As soon as Ql starts to turn on, regenerative feedback via winding P2 will assist the turn-on action bf the transistor, which will now latch to an on state. With Ql on, current will build up linearly in the primary winding at a rate defined by the primary inductance and applied voltage dI dt< xVcc LP). As the current builds up in the collector and emitter of Ql, the voltage across R3 will crease. The...

235 Distribution Of Power Losses

Figure 2.23.3 shows how the power losses are distributed between the two power transistors Ql and Q2 and the series resistor Rl over the output voltage range for the maximum output current of 2 A. Note that the peak power conditions for transistors Ql and Q2 occur at different voltages and that both devices can be mounted on the same heat sink. This need be rated only for the worst-case combination, which never exceeds 41 W. This is considerably lower than the 140 W that would have been...

236 Voltage Control And Current Limit Circuit

Laboratory variable supplies are usually designed to provide constant-voltage or constant-current performance with automatic crossover between the two modes. FIG. 2.23.3 Distribution of power loss in piggyback linear power supply. FIG. 2.23.3 Distribution of power loss in piggyback linear power supply. Figure 2.23.4 shows a typical output characteristic with the supply set for 30 V and 1 A. Load lines for 60 il, 15 tl, and the critical value 30 (i are shown. It will be seen that the mode of...

24 1 Dividual Block Functions

. us elements of the block schematics in Figs. 2.24.4, 2.24.5, and 2.24.6. re considered in more detail. 24. SWnCHMODE VARIABLE PCWER SUPPLIES Figure 2.24.4 shows the internal circuit for the input power section, block 1 the auxiliary supply, block 2 and the power converter, block 3. Block 1, Input Filter. In block 1 the ac line input is taken via the supply switch SW1 and fuse FSI to the input filter inductor LI. Inrush current limiting is provided by thermistor THI in series with the...

241 Step 1 Select Core Size

If a typical secondary efficiency of 85 is assumed (output diode and transformer losses only), then the power transmitted by the transformer would be 130 W. We do not have a simple fundamental equation linking transformer size and power rating. A large number of factors must be considered when making this selection. Of major importance will be the properties of the core material, the shape of the transformer (that is, its ratio of surface area to volume), the...

244 Forced Current Sharing

Current Sharing Power Supply

This method of parallel operation uses a method of automatic output voltage adjustments on each power supply to maintain current sharing in any number of parallel units. This automatic adjustment is obtained in the following way. Because the output resistance in a constant-voltage supply is so low (a few milliohms or less), only a very small output voltage change is required to make large changes in the output current of any unit. With forced current sharing, in principle any number of units...

244 Operating Principles

Consider once again the flyback power section shown in Fig. 2.24.2. Assume energy has been stored in the main transformer T1 during the on period. It will now be shown that this energy may be transferred to the load as any combination of current and voltage, provided that the power conservation criteria are satisfied. The first requirement that must be satisfied (if the core is to continue to operate at a defined flux density under steady-state conditions) is that the equality of the forward...

249 Block Schematic Diagram General Description

Figure 2.24.3 is a block schematic of the basic functional elements of the complete variable switchmode supply. The major functions are described below. Block . Block 1 is the input filter, voltage doubler, and rectification circuit. It converts the 1151230-V ac input to a nominal 300-V DC output to the converter section, block 3. Block 1 also contains current limiting to prevent excessive inrush current when the system is first switched on, and the voltage doubler option for 115-V operation....

274 The Saturable Reactor Power Regulator Application

Consider a reactor wound on a core of ideal square-loop material and fitted in series with output rectifier diode D1 (position A in Fig. 2.21.1). This gives the circuit shown in Fig. 2.21.4. EEG. 2.21.4 Single-winding saturable reactor regulator with simple voltage-controlled reset transistor Ql. EEG. 2.21.4 Single-winding saturable reactor regulator with simple voltage-controlled reset transistor Ql. In the circuit shown in Fig. 2.21.4, assume that the core is unsaturated at a point S3 on the...

32 Selftracking Voltage Clamp

When a transistor in a circuit with an inductive or transformer load is tuned off, the collector will tend to fly to a high voltage as a result of the energy stored in the magnetic field of the inductor or leakage inductance of the transformer. In the flyback converter, the majority of the energy stored in the transformer will be transferred to the secondary during the flyback period. However, because of the leakage inductance, there will still be a tendency for the collector voltage to...

33 Flyback Converter Snubber Networks

The turn-off secondary breakdown stress problem is usually dealt with by snubber networks a typical circuit is shown in Fig. 2.3.2. The design of the snubber network is more fully covered in Part 1, Chap. 18. Snubber networks will be required across the switching transistor in off-line FIG. 2.3.2 Dissipative snubber circuit applied to the collector of an off-line flyback converter. flyback converters to reduce secondary breakdown stress. Also, it is often necessary to snub rectifier diodes to...

37 Example

Consider the parasitic current loop A, B, C. D, and back to A, shown in Figure 1.3.3. Point A is the high-voltage switching transistor package. For a flyback application, the voltage on this transistor may be of the order of 600 Vand the switching frequency typically 30 kHz. Because of the fast switching edges, harmonics-will extend up to several megahertz. Parasitic capacitive coupling (shown as Cpi in the diagram) will exist between the transistor case A and the ground plane B. The tenth...

39 Line Filter Design

The design approach used in Sees. 3.4 through 3.8 was to consider the line filter as an attenuating voltage divider network for common-mode RF noise. This approach is used in preference to normal filter design techniques, asj he source and load impedances are not definable in the powerline environment. The interference noise generator, in switchmode supplies, is very often a high-voltage source in series with a high impedance this tends to a constant-current source. To give good attenuation,...

42 Faraday Screens As Appued To Switching Devices

When components are mounted on heat sinks which are to be thermally linked to the chassis, the normal way of eliminating undesirable capacitive coupling is to place an eHctrostatic screen between the offending component and the heat sink. This screen, normally copper, must be insulated from both the heat sink and the transistor or diode, so that it picks up the capacitively coupled ac currents and returns them to a convenient star point on the input circuit. For the primary components, the star...

44 Faraday Screens On Output Components

For high-voltage outputs, RFI screens may befitted between the output rectifiers and their heat sinks. If the secondary voltages are small, say 12 V or less, the secondary transformer RFI screen and rectifier screens should not be required. The need for Faraday screens on output rectifier diodes can sometimes be eliminated by making the diode heat sink dead to RF voltages by putting the output filter choke in the return line. Typical examples are shown in Fig. 1,4,4a and b. If the diode and...

53 Types Of Fuses

A time-delay fuse will have a relatively massive fuse element, usually of low-melting-point alloy. As a result, these fuses can provide large currents for relatively long periods without rupture. They are widely used for circuits with large inrush currents, such as motors, solenoids, and transformers. These fuses are low-cost and generally of more conventional construction, using copper elements, often in clear glass enclosures. They can handle short-term high-current-transients, and because of...

53 Useful Properties

This type of converter has a number of useful properties that should not be overlooked. First (and particularly important for power FET operation), the voltages on the two power devices cannot exceed the supply voltage by more than two diode drops for any operating condition, provided that fast-action clamping diodes are used for D1 and D2. This very hard voltage clamping action is ideal for power FET operation, as these devices are particularly vulnerable to overvoltage stress. Second, any...

54 Transformer Design

The hard voltage clamping action of the cross-connected primary energy recovery diodes (D1 and D2), and the preference to operate at higher frequency with FET devices, means that the primary and secondary leakage inductances of the transformer will play an important role in the operation of the supply. The energy stored in the primary leakage inductance LLp cannot be transferred to the output circuit it gets returned to the supply. Hence the leakage inductance results in a useless...

610 Resistance Factor R

In Figs. 1.6.4, 1.6.5, and 1.6.6, the effective series resistance R4 has been converted to a resistance factor for more universal application, where If specifications call for a power factor better than 0.6, it may be necessary to supplement the normal sour e resistance with an additional series power resistor. This has a penalty of increased power loss, with an inevitable decrease in overall efficiency. For power factors better than 0.7, a low-frequency chote input filter may be required....

612 Dc Output Voltage And Regulation For Rectifier Capacitor Input Filters

It has been shown 26, 83 that provided that the product to X Cg X RL > 50, the DC output voltage of the rectifier capacitor input filter (with a resistive load) will be defined mainly by the effective series resistance Rs and load power. However, when the ripple voltage is low, this criterion also holds for the nonlinear converter-type load. Figures 1.6.7 and 1.6.8 show the mean DC output voltage of the rectifier capacitive input filter as a function of load power and input m s voltage up to...

614 Selecting Reservoir Andor Filter Capacitor Size

In the above example, the reservoir and or filter capacitor values were chosen to meet the rather simplistic Ce 1.5 aF W criterion indicated in Sec. 6.12. In practice, one or more of the following five major factors may control the selection RMS ripple current rating Ripple voltage Voltage rating Size and cost Holdup time This rating must be satisfied to prevent excessive temperature rise in the capacitor and possible premature failure. (See Part 3, Chap. 12.) The problem at this stage is to...

63 General Operating Principles

In the self-oscillating converters considered here, the switching action is maintained by positive feedback from a winding on the main transformer. The frequency is controlled by a drive clamping action which responds to the increase in magnetization current during the on period. The amplitude at which the primary current is cut off, and hence the input energy, is controlled to maintain the output voltage constant. The frequency is subject to variations caused by changes in the magnetic...

64 Isolated Selfoscillating Flyback Converters

A more practical implementation of the self-oscillating technique is shown in Fig. 2.6.4. In this example, the input and output circuits are isolated, and feedback is provided by an optical coupler OC1. Components D3, C4, and R8form a self-trackingvoltage clamp (see Sec. 3.2). This clamp circuit prevents excessive collector voltage overshoot (which would have been generated by the primary leakage inductance) during the turn-off action of Ql. Components D1 and C3 are the rectifier and storage...

65 Control Circuit Briefdescription

A very simple control circuit is used. The diode of the optical coupler OC1 is in series with a limiting resistor R9 and a shunt regulator U1 (Texas Instruments TL430). When the reference terminal of the shunt regulator VI is taken to 2.5 V, current will start to flow into the cathode of VI via the optocoupler diode, and control action is initiated. The ratio of R12 and Rll is selected for the required output, in this case 12 V. The optocoupler transistor responds to the output control circuit...

66 Rectifier And Capacitor Waveforms

Figure 1.6,3a shows the familiar full-wave rectifier waveforms that would be obtained from the circuit shown in Fig. 1.6.2. The dashed waveform is the half EIG. 1.6.3 Rectifier and capacitor voltage and current waveforms in a full-wave capacitor input filter, (a) Capacitor voltage waveform ( rectifier diode current waveform (c) capacitor current waveform. EIG. 1.6.3 Rectifier and capacitor voltage and current waveforms in a full-wave capacitor input filter, (a) Capacitor voltage waveform (...

67 Summary Of The Major Parameters For Selfoscillating Flyback Converters

The component count is clearly very low, giving good reliability at economic cost. The converter transformer may be designed to operate very near the maximum flux density limit, as the power transistor switches off at a well-defined current level. Any tendency to saturate is recognized by the control circuit, and the on pulse is terminated. (The frequency automatically adjusts to a higher value at which the core will not saturate.) This self-protecting ability leaves the designer free to use...

676 Power Factor And Efficiency Measurements

From Fig. 1.6.3, it can be seen that the input voltage is only slightly distorted by the very nonlinear load presented by the capacitor input filter. The sinusoidal input is maintained because the line input resistance is very low. The input current, however, is very distorted and discontinuous, but superficially would appear to be a part sine wave in phase with the voltage. This leads to a common error The product Vin(rms) x in(rms) is assumed to give input power. This is not so This product...

68 Effective Input Current Ie And Power Factor

In Figs. 1.6.4, 1.6.5, and 1.6.6, the rms input, peak, and ripple currents are all given as a ratio to a calculated effective input current I, FIG. 1.6.4 RMS input current as a function of loading, with source resistance factor R as a parameter. Rsfa son -W I Hsf*1Son-W V BRIDGE Rsf soon -w I Rsf ' 50 il W Rsf 150 -W I VOLTAGE Rsf 500 12 -W f DOUBLER Rsfa son -W I Hsf*1Son-W V BRIDGE Rsf soon -w I Rsf ' 50 il W Rsf 150 -W I VOLTAGE Rsf 500 12 -W f DOUBLER FIG. 1.6.5 RMS filter...

72 Power Limiting And Currentmode Control As Appued To The Selfoscillating Flyback Converter

The self-oscillating complete energy transfer flyback converter responds particularly well to the application of current-mode control. This will be explained by reference to the circuit shown in Fig. 2.6. . The voltage across R4 (which sets the maximum collector current) cannot ex- ceed 0.6 V under any conditions, because at this point Q2 will turn on, turning off the power device Ql. This will occur irrespective of the condition of the voltage control circuit, because the control circuit...

72 Series Resistors

For low-power applications, simple series resistors may be used, as shown in Fig. 1.7.1. However, a compromise must be made, as a high value of resistance, which will give a low inrush current, will also be very dissipative under normal operating conditions. Consequently, a compromise selection must be made between acceptable inrush current and acceptable operating losses. The series resistors must be selected to withstand the initial high voltage and high current stress (which occurs when the...

725 Operating Principles Practical Circuit

A bias voltage is set up on the base of Ql by the current in Rl, Dl, and D2. Q1 conducts to develop a second bias voltage across R2 of approximately one diode drop (0.6 V). The current flow in R3 is similar to that in R2, and a third bias voltage is set up across R3 which is slightly less than that across R2, since the resistance of R3 is lower than that cf R2, Hence, under quiescent conditions, transistor Q2 is close to conducting. At the same time, capacitor C3 will charge through R4,R2,...

73 Thermistor Inrush Limiting

Negative temperature coefficient thermistors (NTC) are often used in the position of Rl, R2, or R3 in low-power applications. The resistance of the NTCs is high when the supply is first switched on, giving them an advantage over normal resistors. They may be selected to give a low inrush current on initial switch-on, and yet, since the resistance will fall when the thermistor self-heats under normal operating conditions, excessive dissipation is avoided. However, a disadvantage also exists with...

73 Voltage Control Loop

Figure 2.7.1 shows the collector and emitter current waveforms of Ql under steady-state voltage-controlled conditions. The emitter cunrent waveform shows a DC offset as a result of the base drive current component Ib. Afi analogue voltage of the emitter current will be developed across R4. The voltage across R5 (Vrj) shows the effect of the snubber current in R5 imposed on the voltage across R4, resulting in a rapid increase in voltage toward the end of the conduction period. The voltage...

7310 Type 3 Overload Protection By Fuses Current Limiting Or Trip Devices

Type 3 employs mechanical or electromechanical current protection devices, and these will normally require operator intervention to be reset. In modem electronic switchmode power supplies, this type of protection is normally used only as a backup to the self-recovery electronic protection methods. Hence it is a last ditch protection method. It is required to operate only if the normal electronic protection fails. In some cases a combination of methods may be used. Included in type 3 protection...

739 Type 2 Output Constant Current Limiting

Power supplies and loads can be very effectively protected by limiting the maximum current allowed to flow under fault conditions. Two types of current limiting are in common use, constant current and foldback current limiting. The first type, constant current limiting, as the name implies, limits the output current to a constant value if the load current tries to exceed a defined maximum. A typical characteristic is shown in Fig. 1.13.1. From this diagram, it can be seen that as the load...

746 Foldback Current Limits In Switchmode Supplies

The previous limitations would also apply to the application of foldback protection in switchmode supplies. However, in switchmode units, the dissipation in the control element is no longer a function of the output voltage and current, and the need for foldback current protection is eliminated. Consequently, foldback protection should not be specified for switching supplies. It is not necessary for protection of the supply and is prone to serious application problems, such as lockout. For this...

75 Using Fieldeffect Transistors In Variablefrequency Flyback Converters

At the time of writing, FETs were available with voltage ratings of up to typically 800 V consequently, their use for flyback off-line converters was somewhat limited. The maximum rectified DC header voltage Vcc for 220-V units, and also for dual input voltage units in which voltage doubler techniques are used, will be approximately 380 V DC. As the flyback voltage stress will normally be at least twice this value, the margin of safety when 800-V power FETs are used is hardly sufficient....

82 Dissipativepassive Start Circuit

Figure 1.8.1 shows a typical dissipative start system. The high-voltage DC supply will be dropped through series resistors R1 and R2 to charge the auxiliary storage capacitor C3. A regulating zener diode ZD1 prevents excessive voltage being developed on C3. The charge on C3 provides the initial auxiliary power to the control and drive circuits when converter action is first established. This normally occurs after the soft-start procedure is completed. The auxiliary supply.is supplemented from a...

82 Operating Principles

Under steady-state conditions, the circuit operates as follows. When transistor Q1 turns on, the supply voltage Vcc is applied to the,primary winding PI, and a secondary voltage Vs will be developed and applied to output rectifier Dl and choke L,. Neglecting diode drops and losses, the voltage across the choke LT will be Vr less the output voltage Vout (assuming the output capacitor Ca is large so that the output voltage can be considered constant). The current in Ls will be increasing linearly...

83 Transistor Active Start Circuit

Figure 1.8.2shows the basic circuit of a more powerful and fast-acting start system, incorporating a high-voltage transistor Ql. In this arrangemiflt, the resistance of R1 and R2 and the gain of Ql are chosen such that transistor Ql will be biased into a fully saturated on state soon after initial switch-on of the supply. As CI and C2 charge, current flows in R1 and R2 to the base of Ql, turning Ql fully on. Zener diode ZD1 will not be conducting initially, as the voltage on C3 and the base of...

831 Minimum Choke Inductance and Critical Load Current

The minimum value of L, is normally controlled by the need to maintain continuous conduction at minimum load current. Figure 2.8.2 shows continuous-mode EE3. 2.8.2 Secondary current waveforms, showing incomplete energy transfer (coninuous-modeoperation) and complete energy transfer (discon-tinuous-mode operation). EE3. 2.8.2 Secondary current waveforms, showing incomplete energy transfer (coninuous-modeoperation) and complete energy transfer (discon-tinuous-mode operation). conduction in the...

84 Impulse Start Circuits

Figure 1.8.3 shows a typical impulse start circuit which operates as follows. Resistors R1 and R2 (normally the discharge resistors for the reservoir capacitors CI and C2) feed current into capacitor C3 after switch-on. The auxiliary supply capacitor C4 will be discharged at this time. The voltage on C3 will increase as it charges until the firing voltage of the diac is reached. The diac will now fire and transfer part of the charge from C3 into C4, the transfer current being limited by...

84 Multiple Outputs

Extra windings on the main transformer can provide additional auxiliary outputs. Once again, the value of secondary voltage will be chosen so that the volt-seconds on the output chokes in the forward and reverse directions will equate to zero under steady-state conditions. Therefore, if the voltage of the main output line is stabilized, the voltage of the auxiliary lines will also be stabilized, provided that the load conditions remain reasonably constant. If the load on any output falls below...

85 Energy Recovery Winding P2

During the on period of transistor Ql, energy will be transferred to the output circuit. At the same time, the primary of the transformer will take a magnetizing current component and store energy in the magnetic field of the core. When Q1 turns off, this stored energy will result in damagingly large flyback voltages on the collector of the switching transistor Q1 unless a clamping, or energy recovery, action is provided. Note Duringthe flyback (off') period, the output diodes will be...

922 Step 2 Select Optimum Induction

The optimum induction Bopt is chosen so as to make the core and copper losses approximately equal. This gives minimum overall loss and maximum efficiency, provided that core saturation is avoided. From Fig. 2.9.1, at 100 W and a frequency of 30 kHz, an optimum peak flux density Bopt of approximately 150 mT is recommended for push-pull operation. Remember, in the push-pull case, the differential excitation (AB) will be twice this peak value, giving a flux density swing of 300 mT p-p. (See Fig....

924 Step 4 Calculate Secondary Turns

The secondary turns will be calculated for the lowest output voltage, in this case 5 V. The output filter and transformer secondary are shown in Fig. 2.93. From Fig. 2.9.3, for the continuous-current-mode operation, FIG. 2.9.3 Output filter of single-ended (buck-derived) I Therefore, the minimum secondary voltage will be 10 V. Allowing for a l-V.drop in diode and inductor, VI becomes 11 V. 1 his must be available at the minimum line input of 90 V and maximum on period of 16.5 p,s. At 90 V in...

925 Multiple Output Applications

Figure 2.9.4 shows a typical multiple-output forward converter secondary, in which all outputs share a common return line. The negative output is developed by reversing D5 and D6. Note that the phasing of the secondary is such that D3 and D5 conduct at the same time during the on period of Ql. Figure 2.9.4 shows a typical multiple-output forward converter secondary, in which all outputs share a common return line. The negative output is developed by reversing D5 and D6. Note that the phasing of...

926 Special Case Half Turns

In this particular example there are two equal and opposite polarity 12-V outputs, and a center-tapped 23-turn secondary winding is used. This arrangement of the positive and negative 12-V outputs is a special case which allows half turns to be used with E cores without causing flux imbalance in the legs of the E core. The mmf from the two half turns (effectively one half on each leg of the E core) will cancel, and the core flux distribution will not be distorted, as shown in Fig. 2.9.5a and b....