102 Operating Principles

FET 1 and FET 2 (the power switches) turn on, or off, simultaneously. When the devices are switched on, the primary supply voltage Vci will be applied across the transformer primary, and the starts of all windings will go positive. Under steady-state conditions, a current will have been established in the output choke LI by previous cycles, and this current will be circulating by flywheel action in the choke LI, capacitor CI, and load, returning via the flywheel diode When the secondary emf is...

103 Overshoot Prevention

The overshoot can be considerably reduced by making the soft-start action very slow, allowing the amplifier to take over before the overshoot is too large. This has the disadvantage that the turn-on delay can be unacceptably long. A much better arrangement is the linear power control circuit shown in Fig. 1.10.3. In this circuit the 2.5-V reference voltage for the control amplifier will be near zero at the noninvertinginput to the amplifier when first switched on, as Cl will be discharged prior...

110 Specifica Tion Notes

The designer should be alert to the tendency for specifications to escalate. When a flyback converter is to be considered and potential requirements are large, costs are often particularly sensitive. The designer should establish with the customer the real limitations of the application. It may well be that a typical performance of 6 regulation on the auxiliary outputs of a multiple-output unit would be acceptable. This allows a semiregulated flyback system to be used. To guarantee a result of...

1101

EES. 1.12.1 Characteristics of a (ypical undervoltage transient protection circuit. (a) Load current transient. (b) Typical undervoltage transient excursion without protection circuit fitted, (c) Undervoltage transient excursion with protection circuit fitted. EES. 1.12.1 Characteristics of a (ypical undervoltage transient protection circuit. (a) Load current transient. (b) Typical undervoltage transient excursion without protection circuit fitted, (c) Undervoltage transient excursion with...

111 Specification Example For A 110w Directoffline Flyback Power Supply

For the following example, a fixed-frequency single-ended bipolar flyback unit with three outputs and a power of 110 W is to be considered. It will be shown later that the same design approach is applicable to variable-frequency self-oscillating units. Although most classical design approaches assume that the mode of operation will be either entirely complete energy transfer (discontinuous mode) or entirely incomplete energy transfer (continuous mode), in practice a system is unlikely to remain...

1113 Transformer Design

The design of the transformer for this power supply is shown in Part 2, Chap. 2. 1.72 PROBLEMS 1. From what family of converters is the flyback converter derived 2. During which phase of operation is the energy transferred to the secondary in a flyback converter . 3. Describe the major advantages of the flyback technique. 4. Describe the major disadvantages of the flyback technique. 5. Why is the transformer utilization factor of a flyback converter often much lower than that of a push-pull...

114 Crowbar Performance

More precise crowbar protection circuits are shown in Fig. 1.11.1b and c. The type of circuit selected depends on the performance required. In the simple crowbar, there is always a compromise choice to be made between ideal fast protection (with its tendency toward nuisance operation) and delayed operation (with its potential for voltage overshoot during the delay period). For optimum protection, a fast-acting, nondelayed overvoltage crowbar is required. This should have an actuation voltage...

115 Limitations Of Simple Crowbar Circuits

The well-known simple crowbar circuit shown in Fig. 1.11. la is popular for many noncritical applications. Although this circuit has the advantages of low cost and circuit simplicity, it has an illdefined operating voltage, which can cause large operating spreads. It is sensitive to component parameters, such as temperature coefficient and tolerance spreads in the zener diode, and variations in the gate-cathode operating voltage of the SCR. Furthermore, the delay time provided by Cl is also...

116 Type 2 Overvoltage Clamping Techniques

FIG. 1.11.2 Shunt regulator-type voltage damp circuits. is highly dissipative, and the source resistance must limit the current to acceptable levels. Hence, shunt clamping action can be used only where the source resistance (under failure conditions) is well specified and large. In many cases shunt protection of this type relies on the action of a separate current or power limiting circuit for its protective performance. An advantage of the clamp technique is that there is no delay in the...

118 Selecting Fuses For Scr Crowbar Overvoltage Protection Circuits

In the event of an overvoltage stress condition caused by the failure of the series regulator in a linear power supply, the crowbar SCR will be required to conduct and clear the stress condition by blowing the series protection fuse. Hence, the designer must be confident that the fuse will open and clear the ulty circuit before the SCR is destroyed by the fault current. If a large amount of energy is dissipated in the junction of the SCR within a short period, the resultant heat cannot be...

119 Type 3 Overvoltage Protection By Voltage Limiting Techniques

In switchmode power supplies, the crowbar or clamp voltage protection techniques tend to be somewhat less favored because of their relatively large size and dissipation. By its nature, the off-line switchmode power supply tends to fail safeM that is, to a zero or low-voltage condition. Most failure modes tend to result in zero output voltage. Since the high-frequency transformer provides galvanic isolation between the input supply and the output lines, the need for crowbar-type overvoltage...

12 Expected Performance

In the example shown in Fig. 2.1.1, the main output is closed-loop-controlled and is thus fully regulated. The auxiliary outputs are only semiregulated and may be expected to provide line and load regulation of the order of 6 . Where better regulation is required, additional secondary regulators will be needed. In flyback supplies, secondary regulators are often linear dissipative types, although switching regulators may be used for higher efficiency. For low-current outputs, the standard...

121 Output Ripple and Noise

Where very low levels of output ripple are required, the addition of a small LC noise filter near the output terminals will often eliminate the need for expensive low-ESR capacitors in the main secondary reservoir positions. For example, a typical 5-V 10-A supply may use the highest-quality low-ESR capacitors in positions Ci, C2, and C3 of the single-stage filter shown in Fig. 2.1.1, but this will rarely give a ripple figure of less than 100 mV. However, it is relatively easy to keep ripple...

1210 Optimum Flux Density

The choice of optimum flux density 5opt will be a matter for careful consideration. Unlike with the flyback converter, both quadrants of the BIH loop will be used, and the available induction excursion is more than double that of the flyback case. Consequently, core losses are to be considered more carefully as these may exceed the copper losses if the full induction excursion is used. For the most efficient design, the copper and core losses should be approximately equal. Figure 2.12.2 shows...

1212 Calcula Ting Primary Turns

Once optimum core size and peak flux density have been selected, the primary turns may be calculated. The transformer must provide full output voltage at minimumline input. Under these conditions, the power pulse will have its maximum width of 16.5 (jls. Hence the minimum primary turns are calculated for this condition. With 90 V rms input to the voltage doubling network, the DC voltage will be approximately 222 V. (See Part 1, Chap. 6.) Consider one half cycle of operation. The capacitors CI...

1215 Control And Drive Circuits

The control and drive circuits used for this type of converter are legion. They range from fully integrated control circuits, available from a number or manufacturers, to the fully discrete designs favored by many power supply engineers. A discussion of suitable drie circuits will be found in Part 1, Chaps. 15 and 16. For reliable operation, the drive and control circuits must provide the following basic functions 1. Soft start. This reduces inrush current and turn-on stress, and helps to...

1216 Flux Doubling Effect

The difference in the operating mode for'the single-ended transformer and the push-pull balanced transformer is not always fully appreciated. For the single-ended forward or flyback converter, only one quadrant of the BIH loop is used, there is a remnant flux B and the remaining range of induction is often quite small. (Figure 2.9,2a and b shows the effect well.) in the push-pull transformer, it is normally assumed that the full BIH loop may be used and that B will be incremented from Bmto each...

122 Operating Principles

Figure 2.12,1 a shows the general arrangement of power sections for the half-bridge push-pull converter. The switching transistors Ql and Q2 form only one side of the bridge-connected circuit, the remaining half being formed by the two capacitors Cl and C2. The major difference between this and the full bridge is that the primary of the transformer will see only half the supply voltage, and hence the current in the winding and switching transistors will be twice that in the full-bridge case....

122 Undervoltage Suppressor Performance Parameters

Figure 1.12. la, b, and c shows the typical current and voltage waveforms-that may be expected at the load end of the DC output lines from a power supply when a large transient load is applied by the load. Figure 1.12.1a shows a large transient load current demand during the period from t, to t,. Figure 1.12.1& shows the undervoltage transient that might typically be expected at the load during this transient. (Assume that the voltage dip is caused by the resistance and inductance of the...

124 Practical Circuit Description

Figure 1.12.4 shows a practical implementation of this technique. In this circuit, switch SW1 or Q1 is replaced by Darlington-connected transistors Q3 and Q4. These transistors operate as a switch and linear regulator. FIG. 1.12.4 Example of an undervoltageprotection circuit. FIG. 1.12.4 Example of an undervoltageprotection circuit. Although Q3 and Q4 are now shown positioned between the two capacitors CI and C2, it has been demonstrated above that since they still form a series network, their...

124 Problem Areas

The designer must guard against a number of possible problems with this type of converter. A major difficulty is staircase saturation of the transformer core. If the average volt-seconds applied to the primary winding for all positive-going pulses is not exactly equal to that for all negative-going pulses, the transformer flux density will increase with each cycle (staircase) into saturation. The same effect will occur if the secondary diode voltages are unbalanced. As storage times and...

125 Currentmode Control And Subharmonic Ripple

A coupling capacitor Cx is often fitted to prevent a DC path through the transformer winding when the supply is linked for 110-V operation. This capacitor is intended to prevent staircase saturation by blocking DC current in the transformer primary. Unfortunately, it can also introduce an undesirable effect, characterized by alternate cycles being high-voltage, narrow pulse width and low-voltage, wide pulse width as a result of the capacitor Cx developing a DC bias. This unbalanced operation...

126 Crossconduction Prevention

Cross conduction can be a major problem in the half-bridge arrangement. Cross conduction occurs when both Ql and Q2 are on at the same instant, usually as a result of excessive storage time in the off'-going transistor. This fault applies a short circuit to the supply lines, usually with disastrous results. Two methods are suggested to stop this effect. The simple approach is to apply a fixed end stop to the drive pulse width so that the conduction angle can never be wide enough to allow cross...

13 Operating Modes

Two modes of operation are clearly identifiable in the flyback converter 1. Complete energy transfer (discontinuous mode), in which all the energy that was stored in the transformer during an energy storage period (on period) is transferred to the output during the flyback period (off period). 2. Incomplete energy transfer (continuous mode), in which a part of the en-ergy scored in the transformer at the end of an on period remains in the transfohner at the beginning of the next on period.

131 Transfer Function

The small-signal transfer functions for these two operating modes are quite different, and they are dealt with separately in this section. In practice, when a wide range of input voltages, output voltages, and load currents is required, the flyback converter will be required to operate (and be stable) in both complete and incomplete energy transfer modes, since both modes will be encountered at some point in the operating range. As a result of the change in transfer function at the point where...

1321 General Conditions

Figure 2.13.1 shows the power section of a typical off-line bridge converter. Diagonal pairs of switching devices are operated simultaneously and in alternate sequence. For example, Ql and Q3 would both be on at the same time, followed by Q2 and Q4. In a pulse-width-controlledsystem, there will be a period when all four devices will be off. It should be noted that when Q2 and Q4 are on, the voltage across the primary winding has been reversed from that when Ql and Q3 were on. In this example, a...

133 Type 1 Overpower Limiting

The first type is a power-limiting protection method, often used in flyback units or supplies with a single output. It is primarily a power supply short-circuit protection technique. This and the methods used in types 2 and 4 are electronic, and depend on the power supply remaining in a serviceable condition. The supply may be designed to shut down or self-reset if the overload is removed. In this type of protection, the power (usually in the primary side of the con- verter transformer) is...

134 Type 1 Forma Primary Overpower Limiting

In this form of power limiting, the primary power is constantly monitored. If the load tries to exceed a defined maximum, the input power is limited to prevent any further increase. Usually, the shape of the output current shutdown characteristic is poorly defined when primary power limiting is used on its own. However, because of its low cost, primary power limiting has become generally accepted in lower-power, low-cost units (particularly in multi-output flyback power supplies). It should be...

135 Type 1 Form B Delayed Overpower Shutdown Protection

One of the most effective overload protection methods for low-power, low-cost supplies is the delayed overpower shutdown technique. This operates in such a way that if the load power exceeds a predetermined maximum for a duration beyond a short defined safe period, the power supply will turn off, and an input power off-on cycle will be required to reset it to normal operation. Not only does this technique give the maximum protection to both power supply and load, but it is also the most...

136 Type 1 Form C Pulsebypulse Overpowercurrent Limiting

This is a particularly useful protection technique that will often be used in addition to any secondary current limit protection. * The input current in the primary switching devices is monitored on a real-time basis. If the current exceeds a defined limit, the on pulse is terminated. With discontinuous flyback units, the peak primary current defines the power, and hence this type of protection becomes a true power limit for such units. With the forward converter, the input power is a function...

142 Foldback Principle

Figure 1.14.1 shows a typical reentrant characteristic, as would be developed measured at the output terminals of a foldback-limited power supply. A purely resistive load will develop a straight load line (for example, the 5-fl load line shown in Fig. 1.14.1). A resistive load line has its point of origin at zero, and the current is proportional to voltage. As a resistive load changes, the straight line (which vill start vertically at zero load i.e., infinite resistance) will swing clockwise...

143 Foldback Circuit Principles As Applied To A Linear Supply

FOLDBACK output CURRENT LIMITING 1.115 14. FOLDBACK output CURRENT LIMITING 1.115 REGULATOR TRANSISTOR AND CURRENT LIMIT CIRCUIT REGULATOR TRANSISTOR AND CURRENT LIMIT CIRCUIT ED3. 1.14.2 (a) Foldback current limit circuit, (b) Regulator dissipation with reentrant protection. ED3. 1.14.2 (a) Foldback current limit circuit, (b) Regulator dissipation with reentrant protection. put voltage is zero (output short circuit). At short circuit, the current in R1 is very small, and the voltage across...

144 Lockout In Foldback Currentlimited Suppues

With the resistive load (the straight-line loads depicted in Figs. 1.14.1 and 1.14.3), there can only be one stable point of operation, defined by the intersection of the NON LINEAR LOAD LINE (LOCK OUT AT 'P2') EK3. 1.14.3 Overload and start-up characteristics of a foldback, current-limited supply, showing performance for linear and nonlinear load lines. NON LINEAR LOAD LINE (LOCK OUT AT 'P2') EK3. 1.14.3 Overload and start-up characteristics of a foldback, current-limited supply, showing...

145 Reentrant Lockout With Crossconnected Loads

Lockout problems can occur even with linear resistive loads when two or more foldback-limited power supplies are connected in series. (This series connection is often used to provide a positive and negative output voltage with respect to a common line.) In some cases series power supplies are used to provide higher output voltages. Figure 1.14,5a shows a series arrangement of foldback-limited supplies. Here, positive and negative 12-V outputs are provided. The normal resistive loads R1 and R2...

15 Energy Storage Phase

The energy storage phase is best understood by considering the action of the basic single-output flyback converter shown in Fig. 2.L2. When transistor Ql is turned on, the start of all windings on the transformer FIG. 2.1.2 Simplified power section of a flyback (buck-boost) converter. will go positive. The output rectifier diode D1 will be reverse-biased and will not conduct therefore current will not flow in the secondary while Q1 is conducting. During this energy storage phase only the...

153 Incorrect Turnoff Drive Waveforms

Surprisingly, it is the application of energetic and rapid reverse base drive during the turn-off edge which is the major cause of secondary breakdown failure of high-voltage transistors with inductive loads. Under aggressive negative turn-off drive conditions, carriers are rapidly removed from the area immediately adjacent to the base connections, reverse-biasing the base-emitter junction in this area. This effectively disconnects the emitter from the remainder of the chip. The relatively...

154 Correct Turnoff Waveform

If the base current is reduced more slowly during the turn-off edge, the base-emitter diode will not be reverse-biased, and transistor action will be maintained throughout turn-off. The emitter will continue to conduct, and earners will continue to be removed from the complete surface of the chip. As a result, all parts of the chip discontinue conducting at the same instant. This gives a much faster turn-off collector-current edge, gives lower dissipation, and eliminates hot spots. However, the...

155 Correct Turnon Waveform

During the turn-on edge, the reverse of the above turn-off action occurs. It is necessary to get as much of the high-resistance region of the collector conducting as quickly as possible. To achieve this, the base current should be large, with a fast-rising edge thus earners are injected into the high-resistance region of the collector as quickly as possible. The turn-on current at the beginning of the on period should be consider ably highe than that necessary to maintain saturation during the...

156 Antisaturation Drive Techniques

FIG. 1.15.1 (a) Base drive current shaping for high-voltage bipolar transistors, (b) Collector voltage, collector current, base drive current, p-a base emitter voltage waveforms. FIG. 1.15.1 (a) Base drive current shaping for high-voltage bipolar transistors, (b) Collector voltage, collector current, base drive current, p-a base emitter voltage waveforms. The base drive current waveforms are shown in Fig. 1.15, b. Although it is not essential to profile the drive current waveform for all types...

157 Optimum Drive Circuit For Highvoltage Transistors

A fully profiled base drive circuit is shown in Fig. 1.15. la, and the associated drive waveforms are shown in Fig. 1.15.16. This drive circuit operate as follows. When the drive input to point A goes positive, current will initially flow via CI and D1 into the base-emitter junction of the switching transistor Ql. The initial current is large, limited only by the source resistance and input resistance to Ql, and Ql will turn on rapidly. As CI charges, the voltage across Rl, R2, C2, and Lb will...

16 Energy Transfer Modes Flyback Phase

When Q1 turns off, the primary current must drop to zero. The transformer ampere-turns cannot change without a corresponding change in the flux density LB. As the change in the flux density is now negative-going, the voltages will reverse on all windings (flyback action). The secondary rectifier diode Dl will conduct, and the magnetizing current will now transfer to the secondary. It will continue to flow from start to finish in the secondary winding. Hence, the set-ondary (flyback) current...

165 Drive Transformer Restoration

For the first part of the on period of the driver transistor Q2, Dl and S2 will be conducting. However, when Ql has turned off and the recovery current in the base-emitter junction of Ql has fallen to zero, S2 and hence Dl will become reversed-biased as a result of the voltage applied to winding P2 via Rl. The start of all windings will now go negative, and current will build up in winding P2, resetting the core back toward negative saturation. At saturation, the current in P2 and Q2 is limited...

166 Widerange Proportional Drive Circuits

Where the range of input voltage and load are very wide, the circuit shown in Fig. 1.16.1 will have some limitations, as follows. When the input voltage is low, the duty-cycle will be large, and Ql may be on for periods considerably exceeding 50 of the total period. Further, if the minimum load is small, LI will be large to maintain continuous conduction in the output filter. Under these conditions, the collector current is small, but the on period is long. During the long on period, a...

169 Proportional Drive With Highvoltage Transistors

IfQl is a high-voltage transistor, it is probable that some shaping of the base drive current will be required for reliable and efficient operation, as shown in Sec, 15.1 of Part 1. Figure 1.16.3 shows a suitable modification to the drive circuit in Figure 1.16.2 for high-voltage transistors base drive shaping has been provided by R4, D3, C2, R3, and Lb. EDS. 1.16.3 Push-pull-type proportional drive circuit with special drive current shap ing for high-voltage transistors. EDS. 1.16.3...

171 Complete Energy Transfer

If the flyback current reaches zero before the next on period of Ql, as shown in Fig. 2.1.5a, the system is operating in a complete energy transfer mode. That EE3. 2.1.5 (a) Primary current waveform I and secondary current waveforms I, (discontinuous-mode) operation, ib) Primary and secondary waveforms for incomplete energy transfer (continuous-mode) operation. EE3. 2.1.5 (a) Primary current waveform I and secondary current waveforms I, (discontinuous-mode) operation, ib) Primary and secondary...

172 I ncom plete Energy Transfer

If, in the circuit example shown in Fig. 2.1.2, the on period is increased and the off' period correspondingly decreased, more energy is stored in he transformer during the on period. For steady-state operation, this extra energy must be extracted during the off' period. If the input and output voltages are to be maintained constant, it will be shown that the load current must be increased to remove the extra energy. The slope of the input and output current characteristics cannot change,...

18 snubber Networks 186 Turnoff Dissipation In Transistor Q1

By the same logic as used above (although the waveform is inverted), CI and transistor Q1 both see the same mean current and voltage during the turn-off period. Hence, the dissipation in the transistor during the turn-off period t, to t, will be the same as the energy stored in Cl at the end of the turn-off period (f2). power dissipated in Q1 during the off period, mW snubber capacitance, (JiF Vcco rating of transistor (70 Vceo is the chosen maximum voltage at lc 0) frequency, kHz

18 Transfer Function Anomaly

Ve effective volume of core and air gap 1. MULTIPLE OUTPUTFLYBACK SWITCHMODE POWER SUPPLIES 2.13 This power is proportional to the shaded area to the left of the BIH curve in Fig. 2.1.6 it is clearly larger for the example in Fig. 2.1.66 (the incomplete energy transfer case). Much of the extra energy is stored in the air gap consequently, the size of the air gap will have a considerable effect upon the transmissible power. Because of the very high reluctance of the air gap, it is quite usual to...

1810 The Wea Ving Lowloss Snubber Diode

As shown above, to reduce secondary breakdown stress during the turn-off of high-voltage bipolar transistors, it is normal practice to use a snubber network. Unfortunately, in normal snubber circuits, a compromise choice must be made between a high-resistance snubber (to ensure a low turn-on current) and a low-resistance snubber (to prevent a race condition at light loads where narrow pulse widths require a low CR time constant). This paradox often results in a barely satisfactory compromise....

182 Snubber Circuit With Load Line Shaping

Figure 1.18. la shows the primary of a conventional single-ended flyback converter circuit PI, Q1 with a leakage inductance energy recovery winding and diode P2, D3. Snubber components Dl, Cl, and R1 are fitted from the collector to the emitter of Ql. Figure 1.18,16 shows the voltage and current waveforms to be expected in this circuit. If load line shaping is required, then the main function of the snubber components is to provide an alternative path for the inductively maintained primary...

183 Operating Principles

During the turn-off edge of Ql, under steady-state conditions, the action of the circuit is as follows. As Q1 starts to turn off at tl (Fig. 1.18.1A), the primary and leakage inductance of Tl will maintain a constant primary current lP in the transformer primary winding. This will cause the voltage on the collector of Ql to rise (tl to t2), and the primary current will be partly diverted into Dl and CI (ls) (Cl being discharged at this time). Hence, as the current in Ql falls, the inductance...

184 Establishing Snubber Component Values By Empirical Methods

Referring again to Fig. 1.18.1a,, unless the turn-off time of Ql is known (for the maximum collector current conditions and selected drive circuit configuration), the optimum choice for Cl will be an empirical one, based upon actual measurements of collector turn-off voltages, currents, and time. The minimum value of Cl should be such as to provide a safe voltage margin between the Vceo rating of the transistor and the actual measured collector voltage at the instant the collector current...

185 Establishing Snubber Component Values By Calculation

Figure 1.18.1b shows typical turn-off waveforms when the snubber network Dl, CI, R1 shown in Fig. 1.18.1 is fitted. In this example, CI was chosen such that the voltage on the collector Vce will be 70 of the VceQ rating of Ql when the collector current has dropped to zero at time f2. Assuming that the primary inductance maintains the primary current constant during the turn-off edge, and assuming a linear decay of collector current in Ql from t, to t2, the snubber current I, will increase...

188 Dissipation In Snubber Resistor

The energy dissipated in the snubber resistor during each cycle is the same as the energy stored in Cl at the end of the off' period. However, the voltage across Cl depends on the type of converter circuit. With complete energy transfer, the voltage on Cl will be the supply voltage Vcc, as all flyback voltages will have fallen to zero before the next on period. With continuous-mode operation, the voltage will be the supply voltage plus the reflected secondary voltage. Having established the...

189 Miller Current Effects

When measuring the turn-off current, the designer should consider the inevitable Miller current that will flow into the collector capacitance during the turn-off edge. This effect is often neglected in discussions of high-voltage transistor ac tion. It results in an apparent collector-current conduction, even when Ql is fully turned off. Its magnitude depends on the rate of change of collector voltage (dVcldt) and collector-to-base depletion capacitance. Further, if the switching transistor Ql...

19 Transformer Throughput Capability

It is sometimes assumed that a transformer operating in the complete energy transfer mode has greater transmissible power than the same transformer operating in an incomplete transfer mode. (It sounds as if it should.) However, this is true only if the core gap remains unchanged. Figure 2,1,6a and b shows how, by using a larger air gap, the same transformer may be made to transfer more power in the incomplete transfer mode than it did previously in the complete transfer mode (even with a...

192 Preventing Cross Conduction

Traditionally, the method used to prevent cross conduction is to provide a dead time (both transistors off), between alternate on drive pulses. This dead time must be of sufficient duration to ensure that the on states of the two power transistors do not overlap under any conditions. Unfortunately, there is a considerable variation in the storage times of apparently similar devices. Also, the storage time is a function of temperature, drive circuit, and collector-current loading. Hence, to...

194 Circuit Operation

Consider Figs. 1.19.2 and 1.19.3 for the initial condition when Q2 is just about to turn on (point tl in the drive waveform). At this instant, input 1 of gate U3 is enabled for an on state of Q2. However, as a result of its storage time, Ql will still be conducting and its collector voltage will be low. Hence, input 2 of U3 will be low. As a result of the gating action of U3, the turn-on of Q2 is delayed until the voltage on the collector of Ql goes high. This does not occur until the end of...

2010 Main Output Inductor Values Buck Regulators

In general, the main inductance LI in the output of a buck regulator filter circuit should be as small as possible to give the best transient response and minimum cost. If a large inductance is used, then the power supply cannot respond rapidly to change5 in load current. At the other extreme, too low an inductance Vflll result in very large ripple currents in the output components and converter circuits which will degrade the efficiency. Further, discontinuous operation will occur at light...

2011 Design Example

Assume that a design is required for the main output inductor LI for a single-ended forward converter and filter, as shown in Fig. 1.20Ja. The specification for the converter is as follows The design approach will assume that the output ripple current must not exceed 30 of I, (6 A p-p in this example). Also, to allow for a range of control, the pulse width at nominal input will be 30 of the total period (that is 10 jxs). To provide an output of 5 Vat a pulse width of 30 , the transformer...

2012 Output Capacitor Value

It is normally assumed that the output capacitor size will be determined by the ripple current and ripple voltage specifications only. However, if a second-stage output filter L2, C2 is used, a much higher ripple voltage could be tolerated at the terminals of Cl without compromising the output ripple specification. Hence if ripple voltage were the only criterion , a much smaller capacitor could be used. For example, assume that the ripple voltage at the terminals of Clean be 500 mV. The current...

202 Basic Requirements

The following section on output-filter design assumes that normal good design practice has already been applied to minimize conducted-mode noise and that RFI filters have been fitted to the input supply lines, as specified in Sec. 3.1. To provide a steady DC output, and reduce ripple and noise, LC low-pass filters (as shown in Fig. 1.20. la) will normally be provided on switching supply outputs. In forward converters, these filters carry out two main functions. The prime requirement is one of...

2021 Type 1 Buck Regulators

Figure 2.20.1a shows the general arrangement of the power stages of a typical buck regulator. Switching device Q1 will be turned on and off by a square-wave drive circuit with a controlled on-to-off ratio (duty ratio control). When Ql is on, the voltage at point A will rise to the supply voltage Vin. Under steady-state conditions, a forward voltage of Vin Vout is n w impressed across the series inductor LI, and the current in this inductor will increase linearly during the period Ql is on. The...

2022 Type 2 Boost Regulators

Figure 2,20.2a shows the general arrangement of the power sections of a boost regulator. The operation is as follows. When Ql turns on, the supply voltage will be impressed across the series inductor LI. Under steady-state conditions, the current in LI will increase linearly in the forward direction. Rectifier D1 will be reverse-biased and not conducting. At the same time (under steady-state conditions), current will be flowing from the output capacitor CI into the load. Hence, CI will be...

203 Control And Drive Circuits

There are many suitable control and drive circuits in both discrete and integrated circuit form. Many of the single-ended control circuits used for the forward and flyback converters can be used for the Cuk regulator. Although many switching regulator control circuits use duty ratio control quite successfully, the more recent current-mode control techniques can be applied. These will yield advantages similar to those found in conventional transformer converters. Research and development work is...

203 Parasitic Effects In Switchmode Output Filters

Figure 1.20.1a shows a single-stage LC output filter (such as might be found in a typical forward converter. It includes the parasitic elements Cc, R ESL, and ESR. The series inductor arm LI shows an ideal inductor L in series with the inevitable winding resistance R. The parasitic distributed interwinding capacitance is included as lumped equivalent capacitor Cc, The shunt capacitor CI includes the effective series inductance ESL and the effective series resistance ESR. The equivalent circuit...

204 Inductor Design For Switching Regulators

It will be clear from the preceding that the inductors (or chokes) play a critical part in the performance of the regulators. These inductors carry a large component of DC current, as well as sustaining a large high-frequency ac stress. The inductor must not saturate for any normal condition, and for good efficiency the winding and core losses must be small. The choice of inductance is usually a compromise. Theoretically, the inductance can have any value. Large values are expensive and lossy,...

205 Highfrequency Choke Example

To get the best performance from the high-frequency choke L2, the interwinding capacitance should be minimized. Figure 1.20.3a shows a 1-in-long ferrite rod choke with a yi6-in diameter, wound with 15 turns of closely packed 17 AWG wire. Figure 1.20,36 shows a plot of phase shift and impedance as a function of frequency for this choke. The phase shift is zero at the self-resonant frequency, which in this case is 4.5 MHz. The impedance plot in Fig. 1.20.3c shows the improvement obtained by...

205 Inductor Design Example

Calculate the inductance required for a 10-A, 5-V type 1 buck regulator operating at 40 kHz with an input voltage from 10 to 30 V, when the ripple current is not to exceed 20 of IDC (2 A). Procedure Maximum ripple current will occur when the input voltage is maximum that is, when the voltage applied across the inductor is maximum. 1. Calculate the on time when the input is 30 V. where tp total period (ton + foff) 2. Select the peak-to-peak ripple current. This is by choice 20 of DC, or 2 A in...

206 General Performance Parameters

Where input-to-output galvanic isolation is not essential, switching regulators can provide extremely efficient voltage conversion and regulation. In multiple-output switchmode power supply applications, independent fully regulated secondary outputs can be provided by these regulators. The performance of the overall power unit can then be extremely good. The user should specify the range of load currents for which full performance is required. This range should be as small as is realistic (the...

206 Resonant Filters

By selecting capacitors such that their self-resonantfrequency is near the switching frequency, the best performance will be obtained. Many of the small, low-ESR electrolytic capacitors have a series self-resonant frequency near the typical operating frequencies of switchmode converters. At the self-resonantfrequency, the parasitic internal inductance of the capacitor resonates with the effective capacitance to form a series resonant circuit. At this frequency, the capacitor impedance tends to...

207 Resonant Filter Example

Figure 1.20.5 shows a typical output stage of a small 30-kHz, 5-V, 10-A flyback converter with a two-stage output filter. (In flyback converters, the transformer inductance and CI form the first stage of the J.C power filter.) A second stage high-frequency filter L2, C2 has been added. For this example, the same 1 in, 5 i6-in-diameter ferrite rod inductor used to obtain plot c in Fig. 1.20.3 is used for L2. The 15 spaced turns on this rod give an inductance of 10 fi.H and a low interwinding...

207 The Ripple Regula

A control technique which tends to be reserved for the buck-type switching regulator is the so-called ripple regulator.17 This is worthy of consideration here, as it provides excellent performance at very low cost. The ripple regulator is best understood by considering the circuit of the buck regulator shown in Fig. 2.20,5a. A high-gain comparator amplifier A1 compares a fraction of the output voltage Vou, with the reference VR when the output fraction is higher than the reference, the series...

208 Commonmode Noise Filters

The discussion so far has been confined to series-mode conducted noise. The f j. ter described will not be effective for common-mode noise, that is, noise voltages appearing between the output lines and the ground plane. The common-mode noise component is caused by capacitive or inductive coupling between the power circuits and the ground plane within the p & supply. Initially this must be reduced to a minimum by correct screening and layout at the design stage. Further reduction of the...

2110 Some Advantages Of The Saturable Reactor Regulator

For low-voltage, high-current secondary outputs, the saturable reactgr control is particularly valuable. The onv-state impedance may be very close to the resistance of the copper winding (afew milliohms in high-current applications). Consequently, the voltage drop across the reactor element will be very low in the on state. In the off state, with the right core material, the inductance and hence the reactance can be very high, and leakage current (magnetization current) is low. Consequently,...

2111 Some Limiting Factors In Saturable Reactor Regulators

The saturable reactor is not a perfect switch. Several obvious limitations, such as maximum and minimum off' and on reactance, have already been mentioned. Some of the less obvious but important limitations will now be considered. I. Parasitic Reset. When the voltage applied to the reactor reverses during the reset period, the main rectifier diode D1 must block (turn off). During this blocking period, there will be a reverse recovery current flowing in the diode. This reverse current flows in...

2112 The Case For Constantvoltage Or Constantcurrent Reset Highfrequency Instability Considerations

At high frequencies the area of the BIH loop increases, giving an increased core loss and a general degradation of the desirable magnetic properties. In particular, some materials show a modification of the BIH loop to a pronounced S-shaped characteristic. This S shape can lead to instability if constant-current resetting is used in the control circuit. This effect is best understood by considering Fig. 2.21.10. If constant-current reset is used, then the magnetizing force H is the controlled...

21142 Step 2 Calculate the Minimum Secondary Voltage Required from the Converter Transformer

The maximum on time is 50 of the total period, or 14.3 jjls at 35 kHz. When the SR is fitted, there will be an unavoidable minimum delay on the leading edge of the on pulse, as a result of the time required to take the core from B, to 5sat, even when the reset current is zero. Previous experience with the 6025 material using fast diodes indicates that this delay will typically be 1.3 (as. (The actual value can be calculated when the turns, core size, and secondary voltage have been...

21143 Step 3 Select Core Size and Turns

In this example, it will be assumed that the transformer secondary has been brought out as a flying lead, and that this wire is to be wound on to the saturable reactor core to form the winding. The core size is to be such that the winding will just fill the center hole. Further, it will be assumed that the wire size on the transformer secondary was selected for a current density of 310 A cm2 and that 10 wires of 19 AWG were used. Assuming a packing factor of 80 , the area required for each turn...

21144 Step 4 Calculate Temperature Rise

The temperature rise depends on the core and winding losses and the effective surface area of the wound core. The core loss of Vitrovac 6025 at 35 kHz and 500 mT is approximately 150 W kg. The weight of the 25 x 15 X 10 core is 17 g, so the core loss is 2.5 W. The copper loss is more difficult to predict, as an allowance must be made for the increase in effective resistance of the wire as a result of skin effect. With a multiple-filament winding of this type, the Fr ratio (ratio of DC...

212 Operating Principles

In simple terms, the saturable reactor is. used in high-frequency switchmode supplies as a flux-saturation-controlled power switch, providing regulation by secondary pulse-width control techniques. The method of operation is best explained by considering the conventional buck regul'*''- circuit shown in Fig. 2.21.1. This figure shows the output LC filter and rectif such as would be found on the secondary of a typical single-ended FIG. 2.21.1 . Typical secondary output rectifier and filter cir-...

213 Simple Power Failure Warning Circuits

Figure 1.21.1 shows a simple optically coupled circuit typical of those often used for power failure warning. However, it will be shown that this type of circuit is suitableonly for type 1 failures, that is, totallinefailureconditions.lt operates as follows. The ac line input is applied to the network R1 and bridge rectifier D1 such that unidirectional current pulses flow in the optical coupler diode. This maintains a pulsating conduction of the optical coupler transistor Ql. While this...

214 Dynamic Power Failure Warning Circuits

The more complex dynamic power failure warning circuits are able to respond to brownout conditions. Many types of circuit are in use, and it may be useful to examine some of the advantages and disadvantages of some of the more common techniques. Figures 1.21.2 and 1.21.3 show two circuits that will ensure that sufficient warning of failure is given for all conditions. In the first example, a fraction of the DC voltage on the power converter reservoir capacitors CI and C2 is compared with a...

215 Independent Power Failure Warning Module

The previous two power failure circuits must be part of the power supply, as they depend on the internal DC header voltage for their operation. Figure 1.21.4 shows a circuit that will operate directly from the line input and is independent of Ihis circuit has its own bridge rectifier D1-D4, which again provides a unidirectional half-sine-wave input to the feed resistor Rl, ZD1, and the optical cou- EE3. 1.21.4 Independent power failure module for direct operation from ac line inputs. EE3....

215 Saturable Reactor Quality Factors

The effectiveness of the saturable reactor as a power switch will be determined by several factors as follows The magnetization current can be considered a leakage current in the off' state of the switch. The reactor's quality as an off' switch that is, its maximum impedance_ will be defined by its maximum inductance. This in turn depends on the permeability of the core in the unsaturated state and the number of turns. Increasing the number of turns will, of course, increase the inductance and...

216 Power Failure Warning In Flyback Converters

Very simple power failure warning circuits can be fitted to flyback converters, because in the forward direction the flyback transformer is a true transformer, providing an isolated and transformed output voltage which is proportional to the applied DC. Figure 1.21.5 shows the power section of a simple single-output flyback supply providing a 5-V output. Diode D1 conducts in the flyback mode of T1 to charge C2 and deliver the required 5-V output. The control circuit adjusts the duty cycle in...

217 Controlling The Saturable Reactor

As explained in Sec. 21.2, to control the saturable reactor in switching regulator applications, it is necessary to reset the core during the off' period to a defined position on the BIH characteristic prior to the next forward power pulse. The reset (volt-seconds), may be applied by a separate control winding (transductor or magnetic amplifier operation see Fig. 2.21.7) or by using the same primary power winding and applying a reset voltage in the opposite direction to the previous power pulse...

217 Fast Power Failure Warning Circuits

The previous systems shown in this section respond quite slowly to brownout conditions, because they are sensing peak or mean voltages. The filter capacitor in the warning circuit introduces a delay. Its value is a compromise, being low enough to prevent a race between the holdup time of the power supply and the time constant of the filter capacitor, but large enough to give acceptable ripple voltage reduction. It is possible to detect the imminent failure of the line before this has fully...

218 Current Limiting The Saturable Reactor Regulator

Figure 2.21.8 shows a simple current-limiting circuit which operates as follows. When the output current is such that the voltage across RI exceeds 0.6 V -< n- FIG. 2.21.8 Saturable reactor buck regulator with current-limiting circuit R1 and Q2. FIG. 2.21.8 Saturable reactor buck regulator with current-limiting circuit R1 and Q2. sistor Q2 will conduct and provide a reset current via D2 when the secondary voltage goes negative, thus limiting the maximum current. When this type of current...

219 Pushpull Saturable Reactor Secondary Power Control Circuit

The discussion so far has been limited to single-ended systems. In such systems, the same time (volt-seconds) is required to reset the core during the off period as was applied to the core to set it during the on period. Therefore, if control is to be maintained under short-circuit conditions, the duty ratio cannot exceed 50 unless a high-voltage reset circuit is provided or a reset tapping point is provided on the SR winding. In the push-pull system shown in Fig. 2.21.9, two saturable reactors...

222 Constantvoltage Suppues

From Fig. 2.22.1, the output characteristics of the constant-voltage supply will be recognized. The normal working range for the constant-voltage supply will be for load resistances from infinity (open circuit) to 1 R. In this range, the load current is 10 A or less. The voltage is maintained constant at 10 V in this working range. At load resistances of less than 1 R, the current-limited area of operation will be entered. In a constant-voltage supply, this is recognized as an overload...

222 Example

Consider the triple-output forward-converter secondary circuit shown in Fig. 1.22.1. Assume that the 5-V output is a closed-loop regulated output, fully stabilized and adjusted. There are two auxiliary 12-V outputs, positive and negative, which are now semiregulated as a result of the closed-loop,control on the 5-V line. Assume that the regulation performance required from the 12-V outputs is such that additional series regulators would aot normally be required (say, 6 ), Further assume that to...

222 The Effect of an Air Gap on the AC Conditions

It is clear from Fig. 2.2.1 b that increasing the core gap results in a decrease in the slope of thpfBi characteristic but does not change the required AEi Hence there is an increase in the magnetizing current iHt This corresponds to an effective reduction in the permeability of the core and a reduced primary inductance. Hence, a core gap does not change the ac flux density requirements or otherwise improve the ac performance of the core. A common misconception is to assume that a core which is...

223 Constantcurrentsuppues

The constant-current supply is not so well known, and therefore the concept can be a little more difficult to grasp. In the constant-current supply, the previous FIG. 2.22.1 Output characteristics of a constant-voltage power supply, showing constant-current and reentrant-current protection locus. FIG. 2.22.1 Output characteristics of a constant-voltage power supply, showing constant-current and reentrant-current protection locus. constant-voltage characteristics are reversed. Figure 2.22.2...

223 Saturable Reactor Voltage Adjustment

Consider the effect of placing a saturable reactor toroid (as described in Part 2, Chap. 21) on the output lines from the transformer to the 12-V rectifiers D1 and D2. These reactors LI and L2 are selected and designed so that they take a time-delay period td to saturate, specified by required VolU X ion U 'on actual Vout 1X5 The extra time delay td is introduced on the leading edge of the output p0wer pulse by the saturable reactor, and the 12.7-V output would be adjusted back to 12 V. This...

223 The Effect of an Air Gap on the DC Conditions

A DC current component in the windings gives rise to a DC magnetizing force Hjx on the horizontal H axis of the BIHloop, ( dc is proportional to the mean DC ampere-turns.) For a defined secondary current loading, the value of DC is defined. Hence, for the DC conditions, B may be considered the dependent variable. It should be noted that the gapped core can support a much larger value of H (DC current) without saturation. Clearly, the higher value of H, HDC2, would be sufficient to saturate the...

23 General Design Considerations

In the following design, the ac and DC conditions applied to the primary are dealt with separately. Using this approach, it will be clear that the applied ac voltage, frequency, area of core, and maximum flux density of the core material control the minimum primary turns, irrespective of core permeability, gap size, DC current, or required inductance. It should be noted that the primary inductance will not be considered as a transformer design parameter in the initial stages. The reason for...

232 60hz Line Transformers

Very often small 60-Hz transformers will be used to develop the required auxiliary power. Although this may be convenient, as it allows the auxiliary circuits to be energized before the main converter, the 60-Hz transformer tends to be rather large, as it must be designed to meet the insulation and creepage requirements of the various safety specifications. Hence, the size, cost, and weight of a 60-Hz auxiliary supply transformer tends to make it less attractive for the smaller switchmode...

2321 Preregulator Operation

The nominal 70-V DC nonregulated header voltage VH is applied to a network of resistor R1 in parallel with transistor Q2, in series with the normal linear regulator transistor Ql, as shown in Fig. 2.23.1. When the power supply has been set to give a low output voltage (for exam- pie, 0 V at 2 A), transistor Q2 is turned off, and the majority of the applied header voltage will appear across Rl. Hence, the maximum dissipation will appear in the resistor, and 01 is relieved of the high-stress...

233 Auxiliary Converters

Very small, lightweight auxiliary power supplies can be made using self-oscillating high-frequency flyback converters. The output windings on the converter can be completely isolated and provide both input and output auxiliary needs, in the same way as the previous 60-Hz transformers. Because auxiliary power requirements are usually very small (5 W or less), extremely small and simple converters can be used. A typical example of a nonregulated auxiliary converter is shown in Fig. 1.23.1. In...